Jia-Chi Samuel Chieh, Satish K. Sharma, Raif Farkouh, Sanghamitro Das, Everly Yeo, Maxwell Kerber, Randall Olsen, Emanuel Merulla, Stephen Badger
IMAGE LICENSED BY INGRAM PUBLISHING
Phased arrays in the last decade have made tremendous progress, mostly as a result of the maturity of silicon beamforming chipsets. With 5G communication links on the horizon, silicon beamformers have been front and center and commercial development has been accelerated, and as a result price economies have scaled [1]. System-level considerations are now being made, specifically regarding the level of integration [2], [3], [4], [5], [6], [7]. There have been many papers in the literature which document various high-performance phased arrays. Some notable published papers on this topic include [8], which presents a dual-band multistandard 5G millimeter-wave phased array; [9], an area-efficient 5G millimeter-wave transceiver front end; [10], a K-band hybrid packaged phased-array receiver with integrated antenna array; [11], a 33.5–37.5 GHz beamforming transceiver with hybrid architecture phase shifters; [12], a wide-band transmit and receive phased array which operates from 16 to 52 GHz; [13], which presents a silicon beamforming chip which can support C-, X-, Ku-, and Ka-bands; [14], a dual-polarized transmit phased array for a common data link; [15], a 1024-element satellite communication receive-only phased array; [16] and [17], which present a wafer-scaled phased array that operates in the G-band; and finally, [18], which presents a D-band submillimeter wave beamforming transceiver in silicon germanium (SiGe). Much of the focus of these papers has been on radio-frequency integrated circuit (RFIC) development, where silicon and SiGe afford the ability to support multiple channels and therefore tighter integration, as well as nuanced design and integration of the beamformer with a printed circuit board (PCB) antenna array. However, only a few papers describe the necessary steps, as well as the equipment needed, to do a line-of-sight (LOS), over-the-air (OTA) microwave link using digitally modulated signals. This is, after all, the final use case for these types of electronically scanned antenna arrays, the ability to close LOS microwave links.
In our previous work [2], we described the design and development of a 64-element transmit and receive (T/R), dual-slant polarized Ka-band phased-array antenna using silicon beamforming chips. Here, we leverage that phased array to demonstrate its OTA data throughput capability. A short distance (0.4 km) data throughput link was tested in [19], where a relatively lower data rate of 30 Mb/s was achieved using a 64-element dual-slant polarized Ku-band phased-array antenna. In this work, we have carried out the data throughput testing with the Ka-band design for a much longer 1.0 km OTA link with data rates as high as 150 Mb/s. The ability to characterize the link gives us insight into the operation of our phased-array antennas; it also allows us to extrapolate the maximum range which can be afforded by these phased-array antennas. We discuss the hardware setup, as well as how OTA testing is performed. “The Phased-Array Antenna Development” section gives details of the silicon beamformer that we used, as well as design insight into the phased-array antenna, which is realized on a multilayer PCB. The “Phased-Array Antenna Characterization” section describes indoor laboratory characterization of the performance of the phased-array. The “Over-The-Air Testing: Outdoor Measurement Results” section describes details of the OTA testing, which was performed at a 1.0-km distance.
Today’s emerging 5G silicon RFICs can enable an extremely low-profile phased-array antenna design. The fully integrated chipset eliminates the need for discrete transceiver blocks and includes a polarization switch, a T/R switch, a low-noise amplifier, a power amplifier, phase shifters, and variable attenuators. The fully integrated chipset reduces overall size, cost, and RF losses. One of the commercial RFICs used for our Ka-band design is the Anokiwave AWMF-0116 [15], whose system-level schematic is shown in Figure 1. This chipset is a single-channel T/R chip that can support dual polarizations and operates between 26 and 30 GHz. The transmit channel has 20 dB of gain, with an OP1dB of 13 dBm. The receive channel exhibits 23 dB of gain, with a noise figure (NF) of 5 dB. Both transmit and receive paths have 6-bit control over both amplitude/attenuators. The chipset also has 6-bit phase shifters (average root-mean-square (RMS) phase error of 5°). This chip uses a double-pole double-throw switch to support dual polarizations. The chip consumes 350 mW on transmit and 250 mW on receive. It should be noted that the AWMF-0116 is no longer recommended for new designs. Table 1 shows a limited sampling of commercially available silicon beamforming chips that operate in the Ka-band. As can be seen, as RF beamformers have progressed, the number of channels a chip can support has increased, reducing the level of complexity required on the phased-array PCB. Additionally, newer generations of RF silicon beamforming chips allow greater output power and lower NFs. This allows phased-array systems to achieve higher effective isotropic radiated power (EIRP) and a lower antenna gain-to-noise temperature (G/T) from a given N-element array.
Figure 1. Single-channel system-level block diagram including the Anokiwave AWMF-0116.
Table 1. Limited sampling of commercial silicon beamforming chips in the Ka-Band.
The top and bottom views of a fabricated prototype of the phased array are shown in Figures 2 and 3, respectively. Figure 4 shows a micrograph image of how the RF beamformers are attached to the PCB. As can be seen, the AWMF-0116 has wafer-level chip-scale packaging (WL-CSP) and a flip-chip attachment is used with fine-pitch solder balls. Since this beamformer supports dual-polarized antennas, we use finite-ground coplanar waveguide (FGCPW) transmission lines with a probe/via antenna feed. The FGCPW has ground vias which are used to improve the isolation between antenna ports. Some significant improvements were made to the phased array presented in this article. The improvements include optimizing the RF transition from the stripline feeder to the 2.92-mm coaxial connector, which was found to have significant mismatch loss in the original design. The pitch between ground stitching vias for the stripline feeder was greatly relaxed to enhance manufacturability and yield. Locking power and serial-peripheral-interface (SPI) connectors were implemented to enhance the resiliency of the interconnects during test events. Most significantly, a 65th RFIC was also added to the beamforming network board (Figure 3) as a preamp (Tx) and postamp (Rx) gain stage to compensate for the loss of the feeder network. This article also expands on details, including transmit characteristics, and reports on utilizing the phased-array antenna in an OTA demonstration. The total size of the PCB, including auxiliary connectors, is 3.5 in × 4.5 in (88.9 mm × 114.3 mm). To support low-cost solutions, a planar antenna integrated onto the circuit board is preferable. Figure 5 shows the multilayer PCB stackup. The antennas reside on a Rogers 4350 substrate, with 10 mils (0.25 mm) between the upper and lower patch elements. The antennas are probe-fed (50 Ω) with two vias with a diameter of 8 mils (0.2 mm), one for the slant +45° polarization, and the other for the slant −45° polarization. To enhance the bandwidth as well as support dual polarizations, a stacked-patch antenna was selected. Readers are encouraged to see articles [26], [27], [28], [29] for details related to the stacked-patch antennas and [30], [31], for circularly polarized variants of this type of antenna topology. The driven patch is 90 mil x 90 mil, and the parasitic patch is 95 mil x 95 mil. Finite array simulations were completed in Ansys HFSS [32] using the domain decomposition method, as shown in Figure 6. This unit cell approach allows for reduced computational times, while accounting for mutual coupling and edge effects. The array has an interelemental spacing of approximately 0.6${\lambda}$ at 28 GHz. The 8x8 array utilizes a stripline corporate feed network with Wilkinson combiners. Embedded NiCr thin-film resistors were utilized because the feed network resides between dielectric layers. Via fencing was implemented on the stripline layer to ensure the proper characteristic impedance of the transmission lines. The simulated excess feeder loss is approximately 14 dB. Figure 7 shows the PCB layout of the bottom layer where the RF beamformers reside, as well as the internal stripline layer. Care was taken to ensure that equal path length matching was achieved.
Figure 2. Top view of the generation 2 Ka-band 64-element T/R phased-array antenna.
Figure 3. Bottom view of the generation 2 Ka-band 64-element T/R phased-array antenna.
Figure 4. AWMF-0116 WL-CSP on the PCB.
Figure 5. PCB stackup (10 mil = 0.25 mm).
Figure 6. Finite-array analysis using HFSS domain decomposition.
Figure 7. Stripline Wilkinson corporate feed network for a 2 × 4 subarray within the PCB.
The AWMF-0116 chipset utilizes a five-wire SPI for applying the beamforming algorithm and controlling the array. A custom LabVIEW graphical user interface was written to control the array along with a National Instruments USB-8452 providing the SPI outputs.
Since the phased-array antenna represents an improvement over the one presented in [2], we performed measurements in a far-field anechoic chamber at San Diego State University’s Antenna and Microwave Lab. Measurements were made from 27 GHz to 29 GHz for both polarizations and for both azimuth and elevation planes. For brevity, azimuth scan patterns from the +45° slant linear polarization and elevation scan patterns from the −45° slant linear polarization are presented. No phase or amplitude calibrations are applied, and as can be seen, the sidelobe levels are nearly equal. For larger arrays, calibration is important to ensure adherence to radiation pattern masks for various standards such as the 3rd Generation Partnership Project. Phased-array calibration is typically done element by element, where a single channel/antenna is turned on, and the amplitude and phase characteristics are measured. Once this is done across the entire array, amplitude and phase offsets can be introduced to ensure minimal phase and amplitude variation across the array aperture. Figure 8 shows the measured scan patterns of the array on the azimuth plane for the +45° slant linear polarization at 27 GHz, 28 GHz, and 29 GHz. As can be seen, the measured cross-polarization levels are well below 15 dB. Figure 9 shows the measured scan patterns of the array on plane for the −45° slant linear polarization, and also shows cross-polarization levels below 15 dB. Methods such as subarray mirrored symmetry [33] can further reduce the cross-polarization levels. The copolarization patterns have symmetric sidelobes, indicating that the standard deviation is tight from chip to chip for both amplitude and phase. The 64-element array can perform without a cold plate and without exceeding temperature limits. With the addition of a cold plate, the temperature can be reduced accordingly.
Figure 8. Slant +45° radiation patterns at (a) 27 GHz, (b) 28 GHz, and (c) 29 GHz.
Figure 9. Slant −45° radiation patterns at (a) 27 GHz, (b) 28 GHz, and (c) 29 GHz.
To test the transmit characteristics, we performed EIRP measurements in our test lab. This measurement is similar to measuring the compression point on a power amplifier, where the input power is slowly swept until the amplifier shows compression or saturation. However, in the EIRP measurement, as the input power is slowly swept from low to high values at the input of the array, the EIRP we are measuring is the combination of the aggregation of the power amplifiers within each of the 64 beamforming chips, as well as the gain of the antenna array. We performed this measurement inside our laboratory, with modular absorber foam walls to mitigate any reflections within our laboratory. A Wiltron 68369B was used as a transmit signal generator set at a frequency of 28 GHz. The signal generator fed the phased array directly, which was set to transmit mode. Approximately 5 ft (1.52 m) across the laboratory we had a Pasternack standard gain horn antenna, which was connected to a down converting mixer and then to the Agilent E4440A spectrum analyzer. This is shown in Figure 10. The phased-array antenna was set to maximum gain. The input signal was slowly swept from −30 dBm to 0 dBm at 28 GHz. Since we know the input power to the array, the path loss from the distance between the Tx and Rx antennas, losses from the test cables and external mixer, as well as the gain of the standard gain horn used on the receive side, we can therefore deduce the EIRP out of our phased-array antenna. Figure 11 shows the measured EIRP compression for this array, as well as a comparison to the array from [2]. Much like in power amplifiers, where the signal or waveform is typically operated at a 3–6 dB back off from compression to preserve linearity, the same can be applied to the phased-array antenna in transmit mode.
Figure 10. Laboratory test setup for measuring EIRP and Tx Compression (5 ft or 1.52 m).
Figure 11. Measured output 1-dB compression point of the phased array at 28 GHz.
As can be seen, the addition of the 65th RFIC compensates for the stripline feeder loss and allows a lower input power to be applied at the input of the array. This is practical because most commercially available radios that operate in the L-band (1–2 GHz) have output powers below 0 dBm. Since the 65th RFIC is placed immediately after the coaxial connector, rather than more strategically in the RF signal path, this 65th RFIC compresses before the 64-element level beamforming chips. This is the reason why the EIRP compresses at around 42.2 dBm. The architecture of the array was completed in this way primarily because of space constraints on the PCB, since the lattice spacing is small in the millimeter-wave frequency regime.
To characterize the performance of our phased-array antenna, we performed an OTA test at San Diego State University. There are very few reports in the literature which demonstrate a millimeter-wave phased array over long distance; rather, most instances are demonstrated over short ranges, such as in [34], which demonstrated a 50 m OTA link. The outdoor range for our test was approximately 1.0 km, and the two test sites were at the same altitudes for easy line of sight. The outdoor range is shown in Figure 12. The receive test setup is shown in Figure 13.
Figure 12. OTA 1.0 km test range at San Diego State University between the Physics Building and the Chapultepec Dormitory.
Figure 13. Test setup on Chapultepec Dormitory, Rx phased-array antenna.
To mimic the phased array and radio configuration in our outdoor testing, we utilized frequency converters. We used DSI Instruments MX30000 for the up/down converters. This mixer comes in a small form factor with an integrated frequency synthesizer which can be programmed through a USB connection. The local oscillator can be tuned from 18 GHz to 30 GHz. The insertion loss of the mixer is measured to be 12 dB at 28 GHz. A Keysight signal generator was used as a proxy for the transmitter, and a spectrum analyzer (Agilent E4440A) was used as a proxy for the receiver. Keysight VSA software was used with the E4440A to recover the constellation diagrams of the digitally modulated signals. Our measurement setup was limited by the instantaneous bandwidth of the signal generator, which was 75 MHz. The transmit frequency was set to 1.8 GHz, which is up-converted to 28 GHz over the air.
Phased-array antennas are used on both sides of the link. Figure 14 shows the recovered constellation of quadrature phase shift keying (QPSK) signals at 2–20 Mb/s, and 8-PSK (phase shift keying) signals at 3–60 Mb/s, with a broadside beam on both the Tx and Rx arrays. The resulting error vector magnitude (EVM) of the digitally modulated signal varied from 8% to 13%. Figure 15 shows the recovered constellations of 16-QAM (quadrature amplitude modulation) and 32-QAM signals from 4 to 60 Mb/s, showing a maximum RMS EVM of 11%. Recovering the digitally modulated signal over various electronically scanned beam angles is one of the advantages of using a phased-array antenna. For this test, the transmit phased array was set to various beam angles from 0° to 60°, in 15° steps. The transmit antenna was physically pivoted to offset the scan angle. Figure 16 shows the recovered constellation over various scan angles (0°, 30°, 45°, and 60°) with a QPSK modulation with a 75-MHz bandwidth. As can be seen, as the beam angle is increased, the EVM also increases. As the beam is scanned up to 45°, the RMS EVM is below 22%. At wide scan angles such as 60°, the gain of the array rolls off, and thus the SNR degrades, causing a higher EVM.
Figure 14. Recovered constellations for various bandwidths and modulations.
Figure 15. Recovered constellations for 16-QAM and 32-QAM.
Figure 16. Recovered constellations for various scan angles.
Table 2 shows the transmit link budget, with an EIRP of approximately 38 dBm, and a 4-dB back-off from the compressed EIRP. This correlates with our measured EIRP from the “Phased-Array Antenna Development” section. Table 2 includes various factors that affect the link budget, such as transmit power, mixer loss, coaxial cable loss, input power, RFIC’s Tx preamp gain block, power divider split loss, excess loss, RFIC Tx-element gain, RFIC output power, spatial combining, and Tx array gain. Similarly, Table 3 shows the Rx link budget, and includes estimated factors such as path loss, polarization misalignment loss, and atmospheric loss. The beamformer chip has an associated NF and, using that, we can also estimate the G/T, as shown in Table 3. Once that is known, we can assume the characteristics of the waveform, which in this case is a QPSK with a 75-MHz bandwidth. We can determine what carrier-to-noise ratio (C/No or CNR), or equivalently the Eb/No which is the CNR per bit, is required given a required bit-error-rate (BER). We estimate that our link margin is approximately 10 dB, which means that we could likely operate these arrays at distances approaching 10 km. To add precision to the link budget estimate, one could also measure the G/T of the phased arrays using the hot and cold method described in [12]. This would add fidelity in the link budget estimate.
Table 2. Tx link budget for QPSK.
Table 3. Rx link budget for QPSK.
We have presented the development and demonstration of a Ka-band T/R phased-array antenna that is capable of maintaining a 1.0 km OTA link, with up to 150-Mb/s data rate at electronically scanned angles up to 60°. We also describe test and evaluation methods for performing pertinent OTA microwave LOS digital communications leveraging these phased-array antennas.
This work was supported in part by the Army C5ISR Center. The authors would like to thank Nhat Truong of San Diego State University for his support during the test.
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Digital Object Identifier 10.1109/MMM.2023.3294878