Moisés Patiño-Gómez, Francisco-Javier Ortega-Gónzalez
©SHUTTERSTOCK.COM/ALEXEY LESIK
Aeronautical radio navigation services (ARNSs) remain one of the main applications of RF technology below 1,500 MHz. Services such as very-high-frequency omnidirectional range (VOR), instrument landing systems (ILSs), distance measuring equipment (DME), and ground-based augmentation systems (GBASs) serve and have served for years to provide aircraft guidance in all phases of flight even in situations of low or zero visibility, contributing decisively to making air travel safe [1], [2], [3].
Currently, economic reasons are leading to the replacement of ground-based ARNSs by satellite-based ARNSs [4], [5]. However, due to issues of vulnerability, the complete elimination of ground-based ARNSs is not envisaged, not even in the modernization plans that are being implemented for the coming years by air transportation agencies [6].
During the past few years, several efforts have been made to modernize ground-based ARNS transmitters, and the application of high-efficiency (HE) amplification techniques [7], [8], [9], [10], [11] is helping to improve their reliability while reducing their maintenance costs. One of the most important practical concerns of ARNS transmitters is their reliability, which strongly depends on their operating temperature and is mainly a direct consequence of the heat power dissipated by their power amplifiers (PAs). The reliability of ARNS transmitters is tested by the harsh working conditions to which they are often subjected in terms of ambient temperature, humidity, and atmospheric pressure. This is because they are sometimes installed in poorly insulated cabins located on airport runways and subjected to extreme temperature variations, in some cases without the possibility of using forced ventilation systems. Hence, an improvement in the efficiency of their PAs can also provide a significant improvement in transmitter reliability, helping to fulfill their technical specifications over their entire operating temperature range. These reliability requirements under tough conditions, combined with the relatively relaxed signal technical specifications of ARNS applications, make these systems a clear candidate to use energy-efficient transmission techniques.
In this article, several energy-efficient transmitters for different ARNS applications and their main technical characteristics are shown. These results are compared to those obtained by the same transmitters using conventional technology, demonstrating that HE transmission technologies can provide important advantages not only in terms of energy but also regarding other technical specifications.
ARNSs basically serve to provide an aircraft with its position information during different phases of its flight. Satellite-based and ground-based systems are the main types of these services.
Satellite-based ARNSs (such as Navstar GPS, global navigation satellite systems, and so on) have been used in civil aviation since 1994, when the U.S. Federal Aviation Administration first authorized the use of GPS [4], [12], [13]. They provide very precise aircraft position information with worldwide coverage [14], [15]. However, these systems are not completely free of problems, with their vulnerability to interferences (unintentional or intentional) being the most important [16]. In addition, other geostrategic considerations must be taken into account, such as the fact that only a few countries have the capacity to deploy and maintain their own satellite-based radio navigation service.
Ground-based ARNSs mainly rely on analog technology [2], and their operating principles have been maintained without major changes since their use became widespread after World War II. Programs such as NextGen, in the United States, or SESAR, in Europe, are focused on phasing out ground-based systems in the coming years in favor of satellite-based systems [17], but due to security and geostrategic reasons, ground-based ARNSs remain a cornerstone in the radio navigation infrastructure, and they are sometimes also used to improve the accuracy of satellite-based systems [5].
There are different types of ground-based radio navigation systems that provide solutions for all phases of air navigation, using infrastructures normally located inside airports and managed by civil or military operators. There is an abundant bibliography that covers, in depth, the different types of these systems and the technology behind them [1], [2], [3], but in a very summarized way, it can be said that the main ground-based radio navigation services are the following:
Figure 1. DVOR and DME systems and signals. FM: frequency modulation; AM: amplitude modulation.
Figure 2. A GBAS system and signal. GNSS: global navigation satellite system; TDMA: time-division multiple access; D8PSK: differential eight-phase shift keying.
All these systems have in common that they use simple analog modulations, mostly double-sideband amplitude modulation (AM) (VOR, ILSs, and DME) or uncomplicated digital modulations, such as differential eight-phase shift keying (D8PSK) (used by GBASs) with a low peak-to-average-power ratio (PAPR). The bandwidths occupied by these services are also narrow (compared to those used by modern digital communications systems), and they range from a few kilohertz (VOR, ILSs, GBASs) up to several hundred kilohertz (DME/TACAN), with peak envelope power (PEP) levels from 100 W (GBASs) to approximately 1 kW (DME).
As mentioned in the previous section, most ARNS systems use analog AM signals or simple digital modulation with relatively narrow bandwidths. The main techniques for the generation of variable envelope signals are the low-level AM and high-level AM architectures [9], [18], [19]. Based on the high-level AM technique, the polar architecture (to amplify both analog and digital signals) is theoretically the most efficient solution in terms of power consumption. The use of Doherty PAs [10], [20], [21], [22] could also be a good idea to enhance the efficiency of low-level modulated transmitters, but considering the low bandwidth and PAPR of ARNS signals, polar amplification is potentially a better solution. Moreover, it is important to mention the outphasing technique [23], which is a less widespread method for HE AM and amplification that is also known as Chireix and linear amplification with nonlinear components. Although some technical aspects still need to be solved for the practical application of this technology, it has great potential, as demonstrated by means of high-quality publications [11], [21], [24], [25].
Traditionally, the architecture used to build ARNS transmitters is low-level AM [Figure 3(a)], in which the carrier wave of the transmitter is amplitude modulated at a low-power stage (known as the modulator), and then, it is linearly amplified by a PA chain until it reaches the power level required by the specific service for which it is intended. Hence, many of the RF power stages of ARNS transmitters are made of linear-mode RFPAs (L-RFPAs) that are based on conventional linear amplification classes (such as class A, class B, or class AB) [8], [9], [26], yielding theoretical peak power efficiencies of 78.5% and maximum average power efficiencies of approximately 50% (in practice, closer to 30% in some cases). This approach is valid only for AM signals, and digital signals require a more complex modulator design.
Figure 3. (a) Low-level and (b) high-level AM topologies. EA: envelope amplifier.
The next most popular technique is known as high-level AM [9], which is shown in Figure 3(b). This technique was originally developed to improve the power efficiency of AM broadcast transmitters over a hundred years ago. In contrast to low-level systems, in this case, the AM of the RF carrier is performed at the transmitter RFPA stages, which vary their supply voltage according to the envelope of the desired AM signal. For high-level AM to be possible, the RFPA needs to operate under saturation conditions so that its output power depends (ideally, in quadratic form) on its supply voltage. Therefore, since both carrier and envelope paths can make use of HE PAs, the resulting system is an HE one.
At its core, the high-level AM technique is nothing more than an application example of a polar modulation architecture [27], [28], but it operates with only one signal coordinate. In other words, the polar architecture makes use of an RFPA working in high-level AM mode, where the drive signal has been phase modulated. The operating principle of the polar mode RFPA (P-RFPA) is to consider a communications signal composed of polar components, an envelope with low-pass characteristics, and a bandpass phase-modulated carrier. These components are amplified separately by different circuits:
Since the amplified signal is finally reconstructed by injecting the amplified envelope into the power supply port of the HE-RFPA (which acts as an amplitude modulator in the P-RFPA architecture), it is necessary that the amplified envelope exhibit the best possible quality in terms of distortion and noise because its defects are transferred to the output, decreasing the quality of the amplified RF signal. Most high-level AM systems, in practice, end up resembling polar systems because the phase distortion produced by the V DD phase modulation conversion of its RFPA must be corrected by the phase modulation of its RF carrier, using a phase polar loop, digital predistortion (DPD), and so on.
The main advantage traditionally attributed to the polar transmission architecture has been the approach’s ability to provide higher power efficiency figures than transmitters that employ conventional linear amplification techniques, but polar transmission also provides other advantages, such as its stability and homogeneous operation at different frequencies and temperatures. Additionally, it provides the possibility to precisely control its output power and, sometimes, the possibility to achieve improvements in the quality of the modulated signal. Figure 4 graphically compares power losses between a linear and a polar transmitter.
Figure 4. Power losses in (a) a conventional (low-level modulated) linear transmitter and (b) a polar transmitter with high-level envelope modulation.
The difficulty of applying the polar mode amplification technique increases with the bandwidth and power level of the transmitted signal, so despite its advantages, in practice, it is not possible to use polar mode amplification for all types of transmitters and situations involving nonconstant envelope signals. This is mainly because the maximum bandwidth that polar systems can manage is mostly limited by the bandwidth of their EAs, and the amplified envelope must meet strict noise and distortion requirements since any defects in this process are directly transferred to the quality of the amplified signal. Therefore, although many valuable research and development efforts have been made to apply this technique to modern equipment, they have shown that the polar technique is not easy to use for communication services combining high-power and wideband signals, such as cellular base stations or TV broadcast transmitters. However, the case of ARNS applications is different since the bandwidth of their signals is moderate, although the power levels handled are high, as is the case for AM broadcasting systems, in which the polar technique has been successfully employed for many years.
Although the use of polar amplification technology for ARNS transmitters is still unusual, it can provide significant technical benefits to these particular applications. The main characteristics and advantages of the polar architecture applied to ARNS transmitters are illustrated in the following sections by showing three HE ARNS P-RFPAs designed, in some cases, to update and replace the conventional linear RFPAs of existing transmitters.
The first example is a power stage designed to increase the efficiency of a DVOR commercial low-level AM transmitter, where its class AB final power stage has been replaced by one based on the envelope elimination and restoration (EER) technique [29], [30], [31], [32], [33], [34], [35]. This power stage can also be used for other services, such as ILSs, with slight modifications. The second example is a prototype of an efficiency-improved RF power stage for a DME polar transmitter, and the third example is an HE GBAS polar transmitter demonstrator. Through these comparisons, it is shown that it is possible to increase the power efficiency and reduce the heat power dissipated by transmitters incorporating these amplifiers in a very significant way without altering performance linearity and even improving the quality of the transmitted signal.
A complete DVOR system combines two transmitters corresponding to its omnidirectional reference signal and its directional variable signal (Figure 1). The reference signal is transmitted by amplitude modulating the system main carrier through a multiplex consisting of three analog signals corresponding to a 30-Hz reference subcarrier, a 1,020-Hz identity subcarrier, and a narrow-band voice signal (with a bandwidth between 300 and 3,000 Hz). Since this is a nonconstant envelope signal, it must be processed by a linear or linearized transmitter. On the other hand, the variable signal of the DVOR service is obtained by frequency modulating a 9,960-Hz subcarrier that is radiated by the sideband directional antenna of the system, and since it is a constant envelope signal, it can be processed by a transmitter made of HE saturated or switch mode RFPAs (class D, class E, and so on) [36], [37]. The P-RFPA in this section has been designed to replace the RF power stage of a DVOR reference signal, which is based on L-RFPAs (a low-level AM architecture) and suffers from power efficiency degradation due to the linearity requirements of its amplification chain.
The proposed P-RFPA topology is based on the EER technique, which has the advantage of making possible the direct upgrade of its low-efficiency L-RFPA without further modifications to the original low-level AM circuits. The application of the EER technique requires additional elements, such as a polar demodulator. In Figure 5, a block diagram of an EER-based P-RFPA shows the complementary stages needed for operation.
Figure 5. The EER-based P-RFPA.
The HE-RFPA for the proposed DVOR P-RFPA is a modified version of a frequency modulation broadcast (88–108 MHz) reference circuit for the MRF1K50H silicon laterally diffused MOS transistor. The passive components of this evaluation board have been modified (both at its input and output matching networks) to make it operate at the low VHF air band from 108 to 118 MHz. The capacitor decoupling network of its power supply port has also been adapted to increase the board’s bandwidth so that it can accept the DVOR envelope signal without distortion. This HE-RFPA is based on a push–pull topology that is very popular at HF and VHF because of its good technical properties and relatively simple implementation. Among the main benefits are the direct combination of the power provided by two transistors, wideband operation, matching network simplification, common source inductance parasitic effects removal, and even harmonics suppression [18], [38], [39]. For comparison between the L-RFPA and the P-RFPA when they are used in DVOR transmitters, this HE-RFPA can also operate in linear mode (as an L-RFPA). A simplified schematic of this RFPA appears in Figure 6. The RFPA can be operated either as an L-RFPA or saturated HE-RFPA (L/HE-RFPA), increasing its drive level. Figure 7 presents its measured efficiency and output power.
Figure 6. A simplified VHF DVOR L-RFPA/saturated HE-RFPA. PCB: printed circuit board.
Figure 7. The measured POUT and ${\eta}_{\text{D}}$ versus the PIN of the VHF DVOR L/HE-RFPA (IDQ = 200 mA).
To implement an HE EA for this DVOR P-RFPA, there are different solutions and architectures that can handle the bandwidth, linearity, and power requirements of the envelope of a DVOR reference signal. Among these design alternatives are switch mode solutions (fast dynamic response power converters and class D audio amplifiers), multilevel-based topologies (class H and class G), and hybrid architectures, where an HE stage and linear amplifier are combined [40], [41], [42], [43].
Since the envelope of the reference signal of the DVOR system has a moderate bandwidth and does not reach zero-voltage levels, a switch mode architecture was considered a suitable choice to design this stage because it can provide the best power efficiency compared to its possible alternatives. The main issue related to switch mode EAs is their inherent noise and spectral spurious signals, but fortunately, the combination of the narrow bandwidth of the DVOR signal with the high switching frequency of the EA allows this issue to be solved by simply adjusting the cutoff frequency of its output low-pass filter (LPF).
The EA used in this EER-based P-RFPA is depicted in Figure 8. It is made of four LMG5200 gallium nitride (GaN) field-effect transistor (FET) half-bridge power stages in parallel that are controlled by a pulsewidth modulation (PWM) signal. GaN power FETs were used in this circuit because they provide higher efficiency, higher switching frequencies, and shorter switching times than their silicon counterparts, which helps to improve the overall efficiency of the EA while rejecting unwanted signal remnants of the PWM process.
Figure 8. The HE switch mode EA for the DVOR P-RFPA. The (a) phase-shift self-oscillating pulsewidth modulation stage, (b) dead time generator, (c) switch mode power stage, and (d) LPF.
Regarding the PWM section of this EA, architectures such as hysteresis-based self-oscillating and phase-shift self-oscillating (PSSO) designs [44], [45] were considered and evaluated, with the PSSO topology being the one finally chosen. Figure 8(d) describes a second-order LPF made of an inductor and a shunt capacitor that restores the original envelope waveform by removing the undesired PWM spectral components. This LPF is matched to the input impedance seen at the power supply port of the HE-RFPA, which acts as a load for the EA [46].
To test the performance of this EA, a synthetic signal was used to simulate the components (reference, identity subcarriers, and voice signal) of the envelope of a DVOR reference signal. This is a three-tone signal consisting of a 30-Hz phase reference tone (m = 30%), a 1,020-Hz identity tone (m = 10%), and a 3,060-Hz tone (m = 30%), which serves to represent the voice component of the multiplex signal.
Figure 9 shows the measured voltage waveform obtained at the output of the EA driven by the test signal described in the preceding. The load at its output is a 1.5-Ω power resistor, which simulates the DVOR HE-RFPA power supply port input impedance. The average power of this signal was P avg = 218 W, and its peak power was PEP = 540 W. The measured average power efficiency of the EA operating in these conditions was ${\eta}_{\text{avg}} = {92}{\%}{.}$
Figure 9. The measured voltage waveform (VOUT) at the EA output port and its drive reference (Vref), driven by the envelope of the DVOR test signal.
The EA board also contains part of a polar demodulator used to extract the envelope and carrier signals from the transmitter input signal. Specifically, it contains an RF power splitter and an envelope demodulator (based on the Analog Devices ADL5511 RF envelope detector).
For the HE-RFPA mentioned in the preceding to be amplitude modulated, a high-power signal must be injected at its input port to make it operate under saturation conditions, so an RF driver stage is needed to provide a constant envelope RF carrier with reasonable efficiency and adjustable output power. This RF driver is based on the Wolfspeed CGH400010F GaN high-electron mobility transistor (HEMT), and the same board also integrates a zero-crossing detector based on the Analog Devices LTC6752 comparator, which is part of the polar demodulator in Figure 5. The output stage of this comparator directly drives the gate of the CGH40010F, providing broadband operation for the input network. The input power limits of this RF driver range from −15 to 24 dBm, which is more than the 25-dB dynamic range required for this system. The RF driver transistor output circuit is made of a fifth-order broadband load network that makes the device operate under soft switching conditions (Figure 10). Figure 11 plots the output power (P OUT) and drain efficiency $({\eta}_{\text{D}})$ versus the supply voltage (V DD) in the low VHF air band.
Figure 10. A simplified DVOR RF driver.
Figure 11. The DVOR RF driver’s measured POUT and ${\eta}_{\text{D}}$ versus its VDD (IDQ = 100 mA).
The complete P-RFPA prototype based on an EER topology (Figure 12) was built by connecting the three printed circuit board assemblies (PCBAs) mentioned before (EA, HE-RFPA, and RF driver). The PCBAs were mounted on a heatsink that did not require forced air cooling to meet the DVOR system temperature requirements.
Figure 12. The DVOR P-RFPA prototype based on the EER technique.
The purpose of replacing a conventional L-RFPA with a P-RFPA is to minimize the heat power losses of the DVOR transmitter power stage in such a way that it would be possible to reduce the operating temperature to improve the mean time to failure (MTTF) while maintaining the quality of the transmitted signal. To know whether these objectives have been effectively achieved and to be as precise as possible, Figure 13 compares the main features of the L-RFPA and the P-RFPA (based on the EER topology). Both systems were tested with the same three-tone signal described before [its probability density function (pdf) is in Figure 13] and at similar output power levels.
Figure 13. The measured ${\eta}_{\text{D}},$ power-added efficiency (PAE), and PLOSS versus the POUT of the (a) DVOR L-RFPA and (b) P-RFPA power amplification stages.
From this figure, the L-RFPA (operated at a 0.5-dB back off) obtains a drain efficiency ${\eta}_{\text{D}} = {57}{\%}$ [similarly, its power-added efficiency (PAE) = 57%] at an average output power P OUT = 185 W, and the heat power losses in these conditions were P LOSS = 140 W. In contrast, the P-RFPA reaches a drain efficiency figure ${\eta}_{\text{D}} = {76}{\%}$ (PAE = 74%) at an average output power P OUT = 215 W, and the heat power losses in this case are P LOSS = 63 W. From these results, it is clear that the proposed polar architecture (P-RFPA) provides not only a significant increase in power efficiency (compared to an L-RFPA) but also a significant reduction in power losses P LOSS. As a consequence, increased transmitter costs resulting from the implementation of this HE architecture would be more than offset by the maintenance cost savings obtained by simplifying the transmitter cooling system.
These improvements in power efficiency and the reduction of the heat power losses would not be useful if the transmitter were not able to meet the linearity requirements of the DVOR service. Figure 14 gives the measured spectra at the output of the L-RFPA (in blue) and P-RFPA (in green) before and after linearization by DPD [47], [48], where it is important to note that the linear mode amplification requires both amplitude and phase DPD, while the polar one requires only its phase component to be linearized by DPD. The quality of the amplified signal by both topologies is enough to meet the linearity requirements of the DVOR service, so it is possible to state that the power efficiency improvements provided by the HE P-RFPA (based on the EER technique) have no cost in terms of linearity and spectral purity.
Figure 14. The three-tone test signal measured spectra at the output of the (a) L-RFPA and (b) P-RFPA for a DVOR signal.
In this section, a P-RFPA demonstrator for the RF power stage of a DME polar transmitter is shown and compared to a conventional L-RFPA architecture. It consists of an HE-RFPA based on a GaN HEMT that is fed by an EA. This prototype implements the low-power mode of the DME service and provides an output power level PEP = 100 W, but it is intended to serve as the basis for the design of a DME transmitter capable of implementing the high-power mode by combining several power stages in a corporate way.
The HE-RFPA of this P-RFPA demonstrator is a broadband suboptimum class E [49], [50] design derived from an L-band radar HE-RFPA (based on the CGHV14250 GaN HEMT), and its operating and design principles are described in [51]. To adapt it, some modifications have been made to its input network. The output network has undergone few changes except for its power supply section, in which some decoupling capacitors have been removed to allow the injection of the modulation envelope signal. Its simplified schematic is in Figure 15, and Figure 16 plots the performance of the dual-mode L/HE-RFPA, as it can be operated in either linear (L-RFPA) or HE mode (HE-RFPA).
Figure 15. A simplified L-Band L/HE-RFPA.
Figure 16. The measured POUT, ${\eta}_{\text{D}},$ and PAE versus the PIN of the DME L/HE-RFPA at 1,000 MHz (VDD = 30 V, and IDQ = 200 mA).
The average efficiency obtained when the L-RFPA was driven by a test signal composed of two Gaussian pulses (Figure 1), using a 10-kHz pulsed-repetition frequency (PRF) and dynamic bias control of the transistor, was ${\eta}_{\text{avg}} = {48}{\%},$ for an output power level PEP = 100 W (Figure 17). Although this power efficiency figure can be considered satisfactory under general conditions, it can be improved if this amplifier is used in a polar architecture. However, given that the duty cycle of a DME transmitter is small, in practice, there are other technical reasons beyond a simple increase in power efficiency that may make the use of a polar architecture in DME transmitters advisable, including output power control and the improvement of the modulation dynamic range.
Figure 17. The measured ${\eta}_{\text{D}},$ PAE, and PLOSS versus the POUT of the (a) L-RFPA and (b) P-RFPA for DME (PRF = 10 kHz).
The EA used for the implementation of the DME P-RFPA is composed of an asynchronous buck converter paralleled to a high-speed power operational amplifier (Figure 18). A current hysteresis loop controls the HE switch mode converter, which acts as a current source to provide the lower-frequency components of the envelope signal [41], [43]. Consequently, the power efficiency of this EA depends on the amplified signal bandwidth, and it is increased when the signal bandwidth is decreased. To build this EA prototype, an ADA4870 high-speed operational amplifier demonstration board and an HE switch mode current source have been combined (Figure 19). Figure 20 shows the measured voltage at the output of the EA for the two-pulse Gaussian test signal as well as the currents measured at different key points of the amplifier.
Figure 18. The simplified EA for the DME P-RFPA.
Figure 19. The DME P-RFPA prototype.
Figure 20. Significant measured voltages and currents on the EA: output voltage (vOUT), output current (iOUT), switch mode source current (iSW), and operational amplifier output current (iop).
The complete DME P-RFPA was built by connecting the L-band HE-RFPA and EA mentioned in the previous paragraphs (Figure 19). In this case (since it is a demonstrator of a DME polar transmitter), the RF drive and envelope signals were supplied directly by laboratory instruments to the L-band HE-RFPA and EA, respectively.
It is necessary to comment that the dynamic range achieved by the high-level AM capabilities of this HE-RFPA is approximately 30 dB. Therefore, to fulfill the pulse mask specification of the DME service, its (originally) constant envelope drive signal (RF carrier phase path) has also been amplitude modulated to avoid nonlinearities at the amplifier output (especially at low levels) due to the transistor feedthrough effect. This technique is known as drive modulation (DM) [52], [53].
As in the case of the DVOR transmitter, a series of comparative measurements highlight the main performance differences between the L-RFPA and the P-RFPA. Figure 17 plots a set of measurements obtained from both systems, driven by the same pulsed signal described before and operated at the same output power level and frequency to provide a balanced and accurate comparison. From these measurements, it can be observed that although the polar transmitter provides some advantage in terms of drain efficiency, which reaches ${\eta}_{\text{D}} = {60}{\%}$ (at PEP = 100 W) compared to ${\eta}_{\text{D}} = {48}{\%}$ achieved by the low-level AM transmitter, in practice, its heat power losses are not significant due to the low duty cycle of a practical DME signal (PAPR = 12.8 dB for PRF = 10 kHz). Therefore, the technical reasons for using the polar architecture in a DME transmitter go beyond increasing the power efficiency to more important aspects, such as circuit and thermal stability, AM dynamic range improvements, precise output power control, and the ease of construction in terms of consistency among different transmitter units.
The use of the DM technique in the polar architecture does not undermine the compliance with the increasingly stringent quality requirements of the DME signal; in contrast, it helps to meet them in a stable and consistent manner. Figure 21 provides the spectrum measured at the output of the L-RFPA and P-RFPA transmitters.
Figure 21. The measured DME pulse spectra at the output of the (a) L-RFPA and (b) P-RFPA.
Regarding DME signal time-domain measurements, Figure 22 shows the pulses of the DME transmitter detected at the output of the L-RFPA and P-RFPA. The contribution of the P-RFPA and the DM techniques to the conformation of the amplified signal can also be observed, where, in red, the shape of the DME pulses modulated (in amplitude) exclusively by the P-RFPA (without the application of the DM technique) shows a clear influence of the drive signal feedthrough. Once the DM technique is applied, the blue and green plots show the pulses detected at the output of the P-RFPA before and after the use of DPD. From Figures 21 and 22, it can be seen that polar mode operation contributes to enhance the dynamic range of the amplified DME signal.
Figure 22. The measured time-domain DME pulses at the output of the (a) L-RFPA and (b) P-RFPA.
The final P-RFPA for ARNSs in this article is intended for GBAS applications. Unlike the rest of the ground-based ARNSs (supported on analog AM implementations), the GBAS service uses D8PSK modulation at a data rate of 10,500 symbols/s.
The HE-RFPA used as the basis for building this GBAS P-RFPA is a push–pull design (Figure 23) in which the balanced structure relies on two coaxial ferrite-loaded baluns [54], [55], [56]. It employs the two-cell CGHV40200PP GaN HEMT because its equivalent output capacitance is relatively low, allowing a reasonable implementation of voltage mode class D operation at VHF frequencies. Its load network results from the combination of a hybrid transformer [57] and two LC resonators. The main role of the hybrid transformer is to allow voltage mode class D operation [58], [59], [60] while providing a noninductive path at the power supply port to inject the amplified envelope. The LC resonators then provide proper load termination for class D operation, harmonic filtering, and dc blocking functions. The HE-RFPA input network can be very simple because of the relatively low gate source capacitance of the transistor. It is made of parallel surface-mounted device resistors, located at the gate terminal of the transistor, that provide an approximately 25-Ω load to every end of its input balun. These resistors are also used for stabilizing and biasing the transistor.
Figure 23. A simplified GBAS VHF L/HE-RFPA.
In addition to its HE mode, this amplifier can also be operated in class AB as an L-RFPA. Figure 24 details the P OUT and ${\eta}_{\text{D}}$ performance versus the input power at different power supply levels of this L/HE-RFPA. In addition, the operation of this amplifier as an L-RFPA was tested using a D8PSK GBAS test signal, whose pdf is shown in Figure 25 (PAPR = 3.4 dB), providing an output power PEP = 100 W and an average efficiency ${\eta}_{\text{avg}} = {52}{\%}$ (at a 1-dB back off). The heat power dissipated by the amplifier under these conditions was P LOSS = 42 W; although this is a reasonably good efficiency figure, it can be significantly improved by making the RFPA operate in polar mode (P-RFPA), as shown in the following.
Figure 24. The measured POUT and ${\eta}_{\text{D}}$ versus the PIN at f = 130 MHz (IDQ = 500 mA) for the GBAS L/HE-RFPA.
Figure 25. The measured ${\eta}_{\text{D}},$ PAE, and PLOSS versus the POUT of the (a) L-RFPA and (b) P-RFPA for the GBAS transmitter.
A GBAS P-RFPA demonstrator was built using the VHF HE-RFPA and, since the power supply requirements for this HE-RFPA were similar to those required by the DME HE-RFPA shown in the preceding, the same EA (with a few changes made to the control hysteresis values and the current source inductor) have been used in this prototype. The envelope and phase-modulated RF carrier required to feed these subsystems were digitally generated by laboratory instruments (no DM was needed to meet the GBAS signal specification). However, in this case, the power efficiency of this EA is higher ${(}{\eta}_{\text{avg}} = {89}{\%}{)}$ because the envelope bandwidth of the GBAS signal is narrower than that of the DME signal. Figure 26 displays the whole GBAS P-RFPA demonstrator.
Figure 26. The GBAS P-RFPA.
The main advantage provided by a P-RFPA when used in a GBAS transmitter lies in its high power efficiency and low heat power losses, which can significantly improve the performance of an L-RFPA, even enhancing the quality of the amplified signal. Figure 25 compares the L-RFPA and P-RFPA power performance for the GBAS service. The average efficiency figure obtained by the P-RFPA demonstrator reaches ${\eta}_{\text{avg}} = {72}{\%}{,}$ and the losses are P LOSS = 18 W (less than half the heat power generated by the L-RFPA) for the same output power PEP = 100 W.
The measured spectra of both amplified signals (at the output of the L-RFPA and P-RFPA) are presented in Figure 27 so that they can be easily compared. From these measurements, it can be seen that the spectrum of the signal amplified by the P-RFPA is not degraded by this architecture and is even better than the spectrum provided by the L-RFPA. The measured error vector magnitude (EVM) performance is described in Table 1, where it can be observed that the EVM figures achieved by the P-RFPA are even slightly better than those provided by the L-RFPA.
Figure 27. Measured spectra of the GBAS D8PSK signal at the output of the (a) L-RFPA and (b) P-RFPA before and after applying DPD.
Table 1. The measured performance of the GBAS transmitters.
ARNS systems are a safety-critical technology that requires high-reliability systems and circuits that have traditionally relied on conservative and tried-and-tested electronic solutions. As a consequence, ARNS transmitters have been built using linear transmitter architectures, such as low-level AM. However, the maturation of HE amplification technologies combined with the technical characteristics of the ARNS signals is allowing the introduction of the polar architecture in ARNS transmitters, which provides several benefits in terms of energy efficiency, stability, and manufacturing consistency. All these improvements are obtained without reducing the quality of the transmitted signals; in contrast, the polar architecture helps to meet the increasingly stringent technical requirements of ARNS transmitters.
The possibilities and advantages of using the polar architecture as an HE amplification technique have been analyzed by showing three polar-based RF power stage demonstrators for ground-based radio navigation transmitters. Then, the performance of the polar modulation technique applied to RF power stages for some of the most popular ARNS (DVOR, DME, and GBASs) was compared against the result provided by conventional class AB amplification techniques.
In the case of the DVOR system, it has been shown that the polar technique can provide a significant reduction in power losses and, consequently, in the heat dissipated by the transmitter and its operation temperature. This contributes to a marked improvement of the MTTF, associated with a reduction of maintenance costs.
The polar amplification architecture applied to a DME system has shown that it is also possible to achieve an improvement of the power efficiency provided by conventional transmitters based on L-RFPAs. However, the main advantage obtained by this topology has been the improvement of the transmitter technical specifications by distributing the modulation function among several stages.
Finally, the demonstrator GBAS polar transmitter achieved a significant power efficiency enhancement over its conventional linear transmitter counterpart while improving the spectral quality of the amplified signal.
All these examples demonstrate that the use of RFPAs as high-level polar modulators fits perfectly with the technical characteristics of ARNS systems, making possible improvements in energy efficiency and linearity figures and providing a very interesting alternative for the systems’ upgrading and modernization.
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Digital Object Identifier 10.1109/MMM.2023.3303590