Peng Zhou, Tingting Zhang, Yongrong Shi
©SHUTTERSTOCK.COM/RAWPIXEL.COM
Today’s fast-growing wireless communication technology has greatly increased the demand for the transmission of high-quality signals with high data rates. Because of its intrinsic advantages of high immunity to external noise, lower electromagnetic interference, harmonic suppression, and better linearity, differential signaling is preferable to single-ended signaling for many applications, including high-speed digital circuits, RF systems, and integrated circuits based on CMOS technology [1] [2], [3], [4]. However, in real-world circuit applications, differential signals are usually accompanied by destructive common-mode (CM) noise, which may be induced by asymmetric differential transmission paths with bending or length mismatch. On the other hand, imbalance of the differential signal channels, including amplitude swings and differences between rising and falling times, could also contribute to CM noise. In addition, the radiation emission caused by CM noise is much stronger than the differential signals, leading to a reduced system reliability [5]. Therefore, CM suppression has been an inevitable challenge for differential circuits and systems.
Up to now, differential architecture has primarily been used to design various microwave devices, such as CM filters (CMFs), differential/balanced filters, and differential antennas. The desired response of these devices is to keep good differential-mode (DM) operation while suppressing CM noise. An overview of differential filters has been reported previously [6]; thus, they are not discussed again here. Instead, in this article, we focus on the development of CM suppression techniques in CMFs and in differential antennas, which are introduced from the perspective of design concepts from reflection to absorption. Examples of several CMFs and differential antennas based on different fabrication processes are then analyzed. Finally, the basic design principles for CM suppression are summarized based on equivalent circuit models.
A CMF is an essential microwave device that is usually used in high-speed digital circuits and systems to suppress CM noise while maintaining the integrity of the differential signals. In practice, owing to inevitable conductor and dielectric losses, ideal all-pass responses cannot be realized for differential signals. Therefore, a good CMF should exhibit 3-dB DM cutoff frequencies that are as high as possible (|Sdd21| < 3 dB). Meanwhile, high-speed differential interfaces, including USB 3.0, peripheral component interconnect express, and High-Definition Multimedia Interface, could support data transmission with data rates of 10 Gb/s, which then poses the design challenge of ultrawideband CM suppression in the frequency domain, especially below 5 GHz. Furthermore, miniaturization is also required for CMFs to be integrated into a high-density circuit layout.
In the past decade, various CMFs with high performance have been proposed based on different types of resonators, including defected ground structures (DGSs), transmission line resonators, and mushroom-shaped multimode resonators. While still in the early stages of research, DGSs are primarily proposed to achieve wideband CM suppression for gigahertz differential signals because of their simple design and low cost [7], [8], [9], [10], [11], [12], [13], [14], [15], [16]. Many representative DGSs have demonstrated effective wideband CM suppression, such as complementary split-ring resonators [7], [8], [9], [10] and dumbbell-shaped [11], U-H-shaped [12], and C-shaped DGSs [13]. For instance, Figure 1 depicts the CMF loaded with periodic dumbbell-shaped DGSs proposed in [11] and the corresponding even-mode equivalent circuit model. Since the symmetric plane between differential lines can be equivalent to the perfect electric boundary for the odd mode, symmetrically loading dumbbell-shaped DGSs have little influence on the quality of the DM transmission. For the even mode, a dumbbell-shaped DGS unit cell is modeled by the parallel connection of an inductor and a capacitor, where ${L}_{\text{DGS}}$ models the total equivalent inductance and ${C}_{\text{DGS}}$ models the total capacitance for the whole dumbbell-shaped DGS unit cell. In most cases, a DGS can be modeled by the parallel connection of an inductor and a capacitor. The desired wideband CM suppression performance can be realized by reshaping the DGSs to tune the quality factor of the LC resonators or by periodically cascading DGS unit cells. However, the applications of CMFs using DGSs are restricted by two main challenges. The parasitic radiation introduced by DGSs would degrade the reliability of circuits and systems. Moreover, CM suppression performance would deteriorate seriously when a metal plane is placed beneath the DGS, which restricts its applications in multilayer environments. The performance degradation of DGSs can be improved by introducing a backed cavity at the cost of complex and multilayer designs [16]. Therefore, a complete ground plane is essential for CMFs for real-world applications.
Figure 1. The CMF using periodic dumbbell-shaped DGSs and corresponding even-mode equivalent circuit model proposed in [11].
To overcome the shortcomings of DGSs, much effort has been devoted to research into multilayer and compact CMFs, which are usually realized using printed circuit board (PCB) or low-temperature cofired ceramic (LTCC) technology [17], [18], [19], [20], [21], [22], [23], [24], [25], [26], [27], [28], [29], [30], [31]. Most of the reported multilayer CMFs are designed based on a quarter-wavelength transmission line resonator [17], [18], [19], [20] or a mushroom-shaped multimode resonator [21], [22], [23], [24], [25], [26]. In [17], a novel CMF is proposed by placing a quarter-wavelength transmission line resonator beneath the differential microstrip lines, and thus CM noise is blocked near the resonant frequency. The CM suppression bandwidth is further broadened by cascading several quarter-wavelength transmission line resonators with different electrical lengths. In [19], we propose a surface-mounted CMF based on LTCC technology, in which differential lines with hybrid transmission line resonators are integrated vertically to achieve simultaneous ultrawideband CM suppression and miniaturization. In [22], a miniaturized CMF consisting of meandered differential lines and a mushroom-like structure is proposed with ultrawideband CM suppression, with a fractional bandwidth of about 100%.
To further enhance the CM suppression bandwidth and reduce the circuit size, we propose a novel surface-mounted ultrawideband CMF based on a quad-mode mushroom-shaped resonator [24], as shown in Figure 2(a). The differential lines on the top layer are designed in a meandered shape to increase capacitance coupling between the differential lines and the mushroom-shaped resonator. Simultaneously, miniaturization can also be achieved with the meandered design. In addition, a pair of shorted stubs is asymmetrically loaded on one side of the mushroom-shaped resonator, which is inspired by an asymmetric short-stub loaded transmission line resonator, to introduce multiple resonant modes. Based on LTCC technology, the proposed CMF is fabricated as a surface-mounted device, which is assembled on the test board for measurement, as shown in Figure 2(b). The measured results shown in Figure 2(c) and (d) demonstrate that the proposed CMF exhibits advantages of high DM cutoff frequency over 10 GHz, four CM transmission zeros, and ultrawideband –10-dB CM suppression from 2.8 to 10.5 GHz with 116% fractional bandwidth.
Figure 2. (a) The proposed surface-mounted CMF using the mushroom-shaped resonator in [24]. (b) Photograph of the test board for measurement. (c) Simulated and measured Sdd21 and Sdd11. (d) Simulated and measured Scc21 and Scc11. Simu.: simulated; Meas.: measured.
Most of the reported multilayer CMFs based on transmission line or mushroom-shaped resonators can be synthesized using the generalized equivalent circuit models in [25], as shown in Figure 3. For the odd mode shown in Figure 3(a), the differential lines can be equivalent to a transmission line with an odd-mode characteristic impedance of 50 Ω owing to the perfect electric boundary at the symmetric plane. For the even mode shown in Figure 3(b), by applying a perfect magnetic boundary to the symmetric plane, the half structure of the proposed CMF can be equivalent to a two-port network, which is formed by the cascaded connection of the transmission line model (${Z}_{1},\,{\theta}_{1}$) and a resonator subnetwork. In the resonator subnetwork, the transmission line (${Z}_{2},\,{\theta}_{2}$), inductor L, and transmission line (${Z}_{2},\,{\theta}_{3}$) are cascaded in sequence. In this circuit model, the inductor L models the total equivalent inductance of the shorted stub and via, and the electrical lengths ${\theta}_{2}$ and ${\theta}_{3}$ are determined by the location where the shorted stub is loaded. The derivations for conditions of multiple CM transmission zeros are presented in [25]. By tuning the meandered design of the differential lines and adjusting the location of the shorted stubs, several CMFs with one [17], two [22], and four CM transmission zeros [24], [25] are proposed to achieve ultrawideband CM suppression.
Figure 3. (a) Odd-mode equivalent circuit model. (b) Even-mode equivalent circuit model.
Electromagnetic bandgap (EBG) technology is another effective technique to achieve CM suppression. In theory, EBGs with quarter wavelengths could be periodically introduced in differential traces to modulate CM characteristic impedances and simultaneously keep uniform DM characteristic impedances without significant discontinuity. In this scenario, a certain stopband is generated for CM signals, while the differential lines are almost transparent to the DM. Compared with CMFs using DGSs and mushroom-shaped multimode resonators, EBGs have some unique advantages. For instance, since no extra resonators are introduced, only two metal layers are needed in EBG-based CMFs. Thus, the ground plane is kept unaltered, which could simplify the CMF design while keeping good signal integrity for the DM. Furthermore, the required level of CM suppression and bandwidth can be predicted with simple approximate analytical expressions. Up to now, various CMFs based on EBGs have been reported, such as differential lines with sinusoidal coupling coefficients [32], uncoupled coplanar waveguide to microstrip transitions [33], and stepped-width grating routing [34]. Figure 4(a) shows the EBG unit cell of a transmission line grating for a CMF design, in which CM characteristic impedances are modulated by narrow (high impedance) and wide (low impedance) quarter-wave sections. The fractional bandwidth of the filter can be calculated using the following equation in [33]: \[{BW} = {2} - \frac{4}{\pi}{\cos}^{{-}{1}}\left(\frac{{Z}_{c,n} - {Z}_{c,w}}{{Z}_{c,n} + {Z}_{c,w}}\right). \tag{1} \]
Figure 4. (a) The transmission line grating routing example in [34]. (b) Notch depth versus ${\Delta}{z}_{c}$ with different low and high ${z}_{c}$ values and the number of cells, ${\Delta}{z}_{c}$ = Zc,n–Zc,w.
In addition, Figure 4(b) shows the CM notch depth and bandwidth versus ${\Delta}{z}_{c}$ and the number of cells. It is clear that more unit cells or a larger difference between high and low CM characteristic impedances could contribute to the deeper notch and larger bandwidth.
To achieve ultracompact circuit size for integration applications, a series of circuit topologies for wideband CMF has been proposed and fabricated as chips based on an integrated passive device (IPD) process in [35], [36], [37], and [38]. A representative circuit topology for a CMF in [38] is displayed in Figure 5(a). Because of the symmetry of this circuit, odd- and even-mode analysis can be utilized to simplify the circuit. For the odd mode, the symmetric plane is equivalent to the perfect electric boundary, and the node P is a virtual short circuit. Therefore, loading CM suppression circuits at the symmetric plane would have only a small influence on the quality of the transmission of differential signals. A corresponding odd-mode equivalent circuit model is shown in Figure 5(b), which supports a Butterworth low-pass response for differential signals. In addition, DM all-pass responses can be realized by the topologies in [37]. For the even mode, the symmetric plane is equivalent to the perfect magnetic boundary. Therefore, the CM suppression performance is determined by the half structure of the proposed circuit topology loaded with a CM suppression circuit. Usually, the end terminal of a CM suppression circuit is a shorted inductor, which can guide the CM currents blocked near the resonant frequencies to flow to the ground plane of the CMF chips.
Figure 5. (a) The circuit topology of the CMF in [38]. (b) The DM equivalent circuit model.
Based on Figure 5(a) and (b), in [38] we propose a novel circuit topology for a CMF to achieve ultrawideband CM suppression and a high DM cutoff frequency. The proposed CMF topology is realized on a gallium arsenide (GaAs) substrate with a thickness of 100 µm, and the GaAs substrate parameters are set as ${\epsilon}_{r}$ = 12.9, and tan ${\delta}$ = 0.006 in ADS Momentum simulations. Figure 6(a) shows the fabricated CMF chip, which is assembled on the PCB test board for measurement, as depicted in Figure 6(b). Figure 6(c) and (d) compares the simulated and measured Sdd21 and Scc21, respectively. It is clear that the measured |Sdd21| can be kept lower than 3 dB from dc to 8 GHz. Meanwhile, the measured Scc21 is kept lower than –10 dB from 2.5 to 8.3 GHz with a fractional bandwidth of 107.4%. The discrepancies between measurements and simulations are caused by the inaccuracy of LC values in fabrications and ADS Momentum simulations since the design kits are not provided by the manufacturer. Fortunately, the measured results show acceptable consistency with the simulations, demonstrating the effectiveness of the proposed CMF topology.
Figure 6. (a) Photograph of the fabricated CMF chip in [38]. (b) A photograph of the test board for measurement. (c) Simulated and measured Sdd21. (d) Simulated and measured Scc21.
All of the aforementioned CM suppression techniques are based on the reflection concept, which would return the reflected CM noise to the front circuits and thus not be eliminated in differential systems. To eliminate CM noise more thoroughly, various CM absorption techniques have also been proposed. In most cases, wideband CM absorption can be realized by adding resistors in the return current paths of the CM noise (at the perfect magnetic boundary at the symmetric plane), which has been demonstrated in absorptive CMFs utilizing DGSs [39], [40], mushroom-shaped multimode resonators [41], [42], [43], and lumped RLC circuit topologies [44], [45]. The basic design principles for absorptive CMFs are similar to reflective ones, which, for clarity’s sake, are not discussed here.
As an example, a novel differential antenna element with wideband CM absorption is provided and analyzed to explain the mechanism of CM absorption based on resistor-loaded networks [46]. Figure 7(a) shows the proposed differential patch antenna element, in which a radiation patch with stepped widths, a ground plane, and the proposed differential feeding network with wideband CM absorption are implemented from the top layer to the bottom layer. The patch on the top layer is designed with stepped widths to operate in dual-mode radiation. A T-shaped coupling aperture is etched on the right side of the middle layer for good impedance match, as shown in Figure 7(b). In addition, the proposed differential feeding network shown in Figure 7(c) is constructed by adding an absorptive stub at the symmetric plane of the balanced microstrip structure. For the odd mode, since the symmetric plane is equivalent to the perfect electric boundary, adding an absorptive stub has only a slight influence on the differential signals. Subsequently, owing to the out-of-phase characteristic, the slot resonant mode of a T-shaped coupling aperture can be excited effectively and further coupled to the radiation patch. For the even mode, the mode mismatch between the CM signals and the T-shaped coupling aperture leads to a few CM signals that can be coupled to the top radiation patch and are mostly absorbed by the proposed differential feeding network. The even-mode equivalent circuit model for the proposed differential feeding network is shown in Figure 7(e), and it is formed by the cascaded connection of the transmission lines (${Z}_{1},\,{\theta}_{1}$) and (${2}{Z}_{2},\,{\theta}_{2}$), the resistor ${2}\,{R}_{1}$, and the shorted stub line (${2}{Z}_{3},\,{\theta}_{3}$).
Figure 7. (a) Proposed differential antenna in [46]. (b) Middle layer: ground plane. (c) Bottom layer: proposed differential feeding network. (d) Side view. (e) Even-mode equivalent circuit model. (f) Measurement setup for the differential antenna. (g) Simulated and measured Sdd11 and realized gains. (h) Simulated and measured Scc11 and corresponding CM absorption ratio.
The measurement setup is displayed in Figure 7(f). The measured results depicted in Figure 7(g) and (h) demonstrate that the proposed differential antenna exhibits advantages of good DM dual-mode radiation, flat realized gains, and a high wideband CM absorption ratio (more than 80%). It is worth noting that there are two CM absorption zeros appearing at 2.67 and 8 GHz, respectively. These two zeros are generated by the quarter-wavelength fundamental mode and high-order resonant mode of the shorted stub (${2}{Z}_{3},\,{\theta}_{3}$). Therefore, a resistor-loaded network to achieve wideband CM absorption has been demonstrated by the proposed differential antenna, while CM suppression in the reported differential antennas is usually based on the reflection concept, which is realized by the mode mismatch between the CM and slot resonant mode [47], [48], [49], [50],[51]. Besides the intrinsic advantages of differential feeding techniques, differential antennas can also be easily integrated with balanced circuits without extra balun, reducing the losses and complexity of systems. In addition, low cross polarization can be achieved because of the out-of-phase characteristic of differential signals, as demonstrated by the proposed differential antenna array in [46].
In this article, recently proposed CM suppression techniques are discussed from the perspective of design concepts from reflection to absorption. Four representative CMFs using DGSs, a mushroom-shaped multimode resonator, an EBG, and a lumped circuit topology are analyzed based on equivalent circuit models to provide design principles for reflective CMFs. Furthermore, a differential patch antenna element with wideband CM absorption is analyzed. A review of the performance comparison of different state-of-the-art techniques is provided in Table 1. It is clear that high DM cutoff frequencies and ultrawideband CM suppression are necessary for practical applications. Obviously, the development of CM suppression techniques is usually accompanied by the evolution of the fabrication process, from PCBs and LTCCs to IPDs. It is foreseeable that more novel CM suppression techniques based on advanced processes will be proposed, many of which will have promising applications in high-speed digital circuits and systems and in CMOS integrated circuits.
Table 1. Performance comparison with related work.
This work was supported by the National Natural Science Foundation of China (61601421, 61971387, and 62031012), the Fundamental Research Funds for the Central Universities under Grant NE2020003, the Research fund projects in advance under Grant 1004-YAH20011, and the Research Funds for Shanghai Academy of Spaceflight Technology under Grant SAST2021-009. The corresponding author is Yongrong Shi.
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Digital Object Identifier 10.1109/MMM.2022.3226551