Yanni Wang, Xuehong Sun, Liping Liu
©SHUTTERSTOCK.COM/DABARTI CGI
With the increasing popularity of fifth-generation (5G) applications, the demand for channel capacity and spectral efficiency for wireless applications has grown exponentially. To meet the increasing data transmission demand, multiplexing techniques based on amplitude, frequency, and polarization are generally used to improve the information transmission [1]. However, such methods are no longer adequate for high-speed and complex signal transmission. As a new wireless communication technology, orbital angular momentum (OAM) technology can effectively increase the system capacity and improve spectrum utilization. A new direction for existing wireless communication multiplexing technology is provided [2].
According to traditional electrodynamic theory, the radiation of electromagnetic (EM) waves carries momentum in addition to energy. Momentum can be divided into linear and angular forms [3]. Linear momentum transmission information is simple, robust, and easy to implement but causes spectrum waste owing to the linear momentum of EM waves being unimodal, with only one independent transmission channel at each carrier frequency. The angular momentum includes spin angular momentum (SAM) and OAM, where SAM is used to describe the polarization state, and OAM is used to describe the spiral phase wavefront. Research on angular momentum was initially conducted in the field of optics, with the SAM of light first being theoretically explored by Poynting in 1909 and experimentally investigated by Beth in 1936 [4]. The experimental results verified that SAM reflects only the polarization state. Letting s be the SAM mode number, then ${s} = \pm{1}$ corresponds to right and left polarization waves, and ${s} = {0}$ corresponds to linear polarization waves. Earlier studies related to the angular momentum of light were almost exclusively concerned with SAM, and the fact that helical phase beams with a phase term $\exp{(}{il}\varphi{)}$ (where j is an imaginary unit, l is the OAM model number, and $\varphi$ is the azimuthal angle) carried OAM was not found until 1992 by Allen et al. [5]. Unlike SAM, OAM is related to spatial distribution rather than polarization. The wavefront of the OAM beam is spirally distributed around the propagation axis and changes by ${2}{\pi}{l}$ after one revolution. Theoretically, OAM can contain an infinite number of orthogonal modes, and the different modes do not interfere with each other. Different OAM modes can be employed for encoding information to increase channel capacity, both in the free-space [6] and fiber-optic [7] transmission domains. In addition to OAM having the problem of intermodal intercode crosstalk in practical applications, OAM beams are less robust during propagation and susceptible to interference during transmission [8]. However, OAM still provides a new direction for EM wave transmission beyond the existing EM wave properties. Therefore, OAM can be used as a new coding method for information coding in optical communication applications, and it can realize wavelength division multiplexing, time division multiplexing, polarization multiplexing, and other mode division multiplexing (MDM) for optical communication to extend the capacity of optical communication [9]. In addition, the channel capacity and spectral efficiency of optical communication systems can be improved by combining with other multiplexing methods [10]. In parallel, the optical vortex of OAM beams in superresolution imaging can be used to improve image resolution, image fidelity, and other parameters [11], while the rotational Doppler effect of OAM beams can be used to detect rotating objects [12]. OAM is also applied in optical tweezers to detect particle rotation [13]. In quantum mechanics, at the single-photon level, OAM modes can be used for high-dimensional entanglement. To summarize, OAM has received significant attention in multidisciplinary research fields.
Despite such attention, OAM had not been applied to microwave RF communications until 2007. Thidé et al. later identified that phased array antennas could generate OAM vortex EM waves, which acted as a foundation for applying OAM in microwave RF communications [14]. The application of OAM to wireless communication enables OAM signals of different modes to share the same carrier frequency for transmission, and the receiver can be used orthogonality to perform low-crosstalk separation of the received multimodal OAM waves. The spectrum utilization can be effectively improved by OAM-based multiplexing technology without relying on traditional resources such as time and frequency. As such, designing a simple and compact OAM antenna structure that is easy to process is of considerable practical significance. At present, there are various existing methods that have been proposed by domestic and international scholars for generating high-performance OAM beams. Examples of such methods include spiral paraboloidal structures, spiral phase plate (SPP) structures, metasurface antennas, patch antennas, and waveguide resonant antennas.
China’s Ministry of Industry and Information Technology has conducted technology development trials in the millimeter-wave (mm-wave) band of 5G mobile communication using 30 GHz to 300 GHz [15]. mm-wave communication enables high-capacity information transmission and effectively reduces the antenna size [16]. In the rapidly developing information era, the development of mm-wave communication technology will effectively alleviate the problem of spectrum resource constraints [17]. The mm-wave band is particularly critical in meeting high traffic demands and ensuring the quality of communication systems due to the abundant spectrum bandwidth resources. The industry has shown strong demand for the mm-wave band, and major equipment manufacturers have continued to make breakthroughs in mm-wave research. With the unique advantages thereof, the mm-wave band will undoubtedly have broad application prospects. The combination of the mm-wave band and OAM will further improve the spectrum utilization and expand the system capacity. However, so far, most of the review-type articles on OAM generation and detection methods remain in the C-band, X-band, or Ku-band. There is no article in which OAM generation and detection methods on the mm-wave band are comprehensively summarized. Therefore, in the present article, a comprehensive review of OAM generation and detection methods over the mm-wave band is provided, as well as relevant primary research and the latest developments in terms of technical progress.
In the present article, the transmission characteristics of mm waves and the basic concepts and properties of OAM are described in the “Principle” section. In the sections “Generation Methods of OAM in the mm-Wave Band” and “Detection Methods of OAM in the mm-Wave Band,” the OAM beam production and detection methods in the mm-wave region are discussed, as well as the advantages and disadvantages of such techniques. In the section “Applications of mm-Wave OAM in Communication,” the application of OAM in the mm-wave band is explored. Finally, a summary and perspectives are provided in the section “Conclusion and Future Work.”
The extension of multiple antenna resources and the development of high-bandwidth communication technologies are two common approaches adopted to meet the communication requirements of high-density access users in wireless communication networks. Among the two, the aim of the expansion of multiple antenna resources is to directly or indirectly increase the antenna gain during transmission and obtain higher throughput in the spatial dimension, so as to meet the requirements of multiconnected devices. At the same time, the sizeable supporting number of RF link modules poses a considerably large burden with respect to energy efficiency. Therefore, developing high-bandwidth information transmission methods such as those incorporating the mm-wave band is vital for overcoming the current problem of spectrum resource constraints. The transmission characteristics of the mm wave are described as follows.
In free-space transmission, the ratio of the received signal power to the transmitted power satisfies the Friis transmission equation: \[\frac{{P}_{r}}{{P}_{t}} = {G}_{t}{G}_{r}{\left({\frac{\lambda}{{4}{\pi}{d}}}\right)}^{2} \tag{1} \] where ${\lambda}$ is the wavelength of the spatial EM wave, ${P}_{t}$ is the output power at the transmitter, ${P}_{r}$ is the received power at the receiver, ${G}_{t}$ is the antenna gain at the transmitter, ${G}_{r}$ is the antenna gain at the receiver, and d is the distance between the transmitting and receiving antennas. From the analysis of the Friis transmission equation, if all conditions, including antenna gain, are the same, the loss of each path is inversely proportional to the square of the wavelength. Here an increase in the frequency of the EM wave signal will decrease the received power. The path loss of mm waves is much higher than that of low-frequency signals [18], and the most common statistical model for describing the average path loss (excluding small-scale fading) by linearity is as follows: \[{\text{PL}}{(}{d}{)[}{\text{dB}}{]} = {\alpha} + {10}{\beta}\,{\log}_{10}{(}{d}{)} + {\xi},{\xi}\,{\in}\,{N}\left({{0},{\sigma}^{2}}\right) \tag{2} \] where ${\alpha}$ and ${\beta}$ are linear model parameters, d is the distance, and ${\xi}$ is the log-normal term. The Friis transmission equation is a particular case of ${\beta} = {2}$ in (2). As well as the path loss problem, another major challenge for mm-wave communication is the considerably high attenuation effect of obstacles on mm waves. For instance, brick walls can attenuate mm waves by as much as 40 to 80 dB, and the human body can cause a 20- to 35-dB loss of EM waves in the mm-wave band. Tree foliage occlusion can cause severe attenuation, while the absorption of mm waves by oxygen and rainfall is significant; that is, the large-scale fading of the channel is more severe compared with that of the low-frequency channel. As such, different application scenarios should be considered in the channel model construction of mm waves, including residential buildings, office buildings, classrooms, shopping malls, and others. Meanwhile, weather effects such as rain, snow, and cloudy days should also be considered.
Despite having these shortcomings, mm-wave communication also has several significant advantages:
The basic theory of EM fields states that EM waves carry both energy and momentum as they propagate through space. The momentum can be divided into linear momentum and angular momentum. Angular momentum can be expressed as follows: \[{J} = {p}\,{\times}\,{r} \tag{3} \] where p is the linear momentum and r is the vector radius of the electric field. The SAM is related to the polarization of the EM wave and can be expressed as \[{S} = {\varepsilon}_{0}\int{{\text{Re}}^{\ast}}\,{\times}\,{A}{dV} \tag{4} \] where ${\varepsilon}_{0}$ is the dielectric constant, E is the electric field strength, and A is the vectorial potential function. The OAM is related to the distribution of the EM field and can be expressed as \[{L} = {\varepsilon}_{0}\int{\text{Re}}\left\{{{iE}^{\ast}{[}{-}{i}{(}{r}\,{\times}\,\nabla{)}\,{\cdot}\,{A}{]}}\right\}{dV} \tag{5} \] where i is an imaginary unit. In summary, the angular momentum J can be expressed as \[{J} = {S} + {L} \tag{6} \]
Generally, the values of SAM are $ + {1}$, 0, and ${-}{1}$, which indicate the different polarization modes of EM waves. When the value of SAM is $ + {1}$, the EM waves exhibit right-handed polarization; when the value is ${-}{1}$, the EM waves exhibit left-handed polarization; and when the value of SAM is 0, the EM waves exhibit line polarization characteristics.
OAM can also be understood as being formed by adding a phase factor ${e}^{{il}\varphi}$ to the usual EM wave, which can be expressed as \[{\text{U}}{(}{\text{r}},\varphi{)} = {\text{A}}{(}{\text{r}}{)}{e}^{{il}\varphi} \tag{7} \] where A(r) is the amplitude of the EM wave, r denotes the radiation distance from the beam’s central axis, $\varphi$ is the azimuth angle, and l denotes the mode of the EM wave. When ${l} = {0}$, the wave behaves as a planar EM wave, and when l takes other nonzero integers, the phase distribution diagram of the EM wave presents a vortex shape, exhibiting a spiral phase structure. As shown in Figure 1, the spatial spiral phase distributions are shown for modes ${l} = {0}$, ${l} = \pm{1}$, ${l} = \pm{2}$, ${l} = \pm{3}$, and ${l} = \pm{4}$, respectively. When the phase changes by ${2}{\pi}$, the model is defined as 1; when the phase changes by ${4}{\pi}$, the model is defined as 2. The EM wave’s phase carrying OAM l will change by ${2}{\pi}{l}$. When the value of the OAM mode is positive, a vortex phase wavefront with counterclockwise rotation is obtained. When the value of OAM mode is negative, a vortex phase wavefront with clockwise rotation is obtained. Table 1 shows the properties, principles, and corresponding applications of EM vortex waves.
Figure 1. The OAM phase distributions for different modes. (a) l = 1, (b) l = −1, (c) l = 2, (d) l = −2, (e) l = 3, (f) l = −3, (g) l = 4, and (h) l = −4.
Table 1. The properties, principles, and corresponding applications of EM vortex waves.
Unlike conventional wireless communication technology, the multiplexing principle of the OAM system is that the number of modes of OAM waves is used as a modulation parameter by exploiting the orthogonality among OAM beams of different modes. The information is loaded onto the OAM waves of different modes at the transmitter side, transmitted through the same frequency channel, and demodulated at the receiver side to return the information carried by the OAM beams of different modes. In OAM wireless communication systems, the communication system’s capacity can be increased by simultaneous cofrequency multiplexing of a set of orthogonal OAM modes. Theoretically, by increasing the number of OAM modes, the system’s data transmission rate and spectrum utilization can be increased. An OAM–MDM system was proposed by Zhang et al. in 2016 [19], as shown in Figure 2. Four independent baseband signals are generated by software-defined radios (SDRs) using orthogonal frequency division multiplexing. They are fed to each of the four intermediate frequency ports and then fed to the local oscillator to achieve 10 GHz carrier power. Then the four sets of signals are fed to the four excitation ports through power amplifiers and bandpass filters and modulated to the four independent channels. After near 10 m free-space transmission, the four coaxially propagating data streams are demultiplexed by the receiver antenna. Amplified by the low-noise amplifiers and down-converted to the intermediate frequency signals, signals are fed to the SDR platform where a second amplifying and down-conversion is performed to get the digital baseband signals. Finally, the four OAM modes are multiplexed and demultiplexed at 10 GHz. The theoretical analysis and experimental results show that the spectral efficiency of the system was improved by a factor of 4 compared with the conventional line-of-sight (LOS) multiple input/multiple output (MIMO) system, while the reception complexity was reduced.
Figure 2. (a) The schematic diagram of the 4 × 4 OAM–MDM link. (b) The outdoor experimental setup of the communication link. (Source: Zhang et al. [19]; used with permission.)
A schematic of the OAM beam multiplexing system is shown in Figure 3. Each mode can be divided into an x-direction component and a y-direction component according to the polarization direction, and the signal propagates along the z-axis direction. The vortex wave expression is ${U}\left({{r},\varphi}\right) = {A}\left({r}\right){e}^{{il}\varphi}$ where r is the distance between the far-field observation point and the axial center of the beam, assuming that the time-domain expression of the transmitted signal is S(t); thus, the expression of the tuned signal is \[{U}_{i}{(}{r},\varphi,{t}{)} = {S}{(}{t}{)}{A}{(}{r}{)}\exp{(}{il}\varphi{)} \tag{8} \]
Figure 3. The schematic diagram of OAM beam multiplexing.
Assuming that N signals need to be transmitted, in the vortex wave multiplexing technique, each signal requires a different mode of vortex EM wave as the carrier. The nth tuned signal can be expressed as follows: \[{U}_{sn}\left({{r},\varphi,{t}}\right) = {S}_{n}\left({t}\right){A}_{n}\left({r}\right)\exp\left({{il}_{n}\varphi}\right) \tag{9} \]
The total tuned signal can be obtained by superimposing the N-channel signal, which can be expressed as \[{U}_{s}\left({{r},\varphi,{t}}\right) = \mathop{\sum}\limits_{{n} = {1}}\limits^{N}{{S}_{n}\left({t}\right){A}_{n}\left({r}\right)\exp\left({{il}_{n}\varphi}\right)} \tag{10} \]
For the receiver side, the vortex EM wave has specific anti-interference capabilities due to the orthogonality between the modes, and thus, the received tuned signal is \[{U}_{s}^{\text{Re}}\left({{r},\varphi,{t}}\right) = \mathop{\sum}\limits_{{n} = {1}}\limits^{N}{{S}_{n}\left({t}\right){A}_{n}^{\text{Re}}\left({r}\right)\exp\left({{il}_{n}\varphi}\right)} \tag{11} \]
For the received tuned signals, each signal in both directions needs to be demodulated. Here the assumption is that the demodulation is performed for the pth signal. Usually, the vortex EM waves of opposite modes are needed for demultiplexing, and for the pth way tuned signal, a phase factor of ${e}^{{-}{il}_{p}\varphi}$ needs to be superimposed on the tuned signal.
In vortex EM wave multiplexing, each signal is modulated with a different modal EM wave, and during the modulation process, an additional EM wave with opposite modalities is required for each signal. Such a multiplexing technique can transmit multiple signals simultaneously and at the same frequency, thereby providing a theoretical basis for improving the channel capacity and spectrum utilization.
Currently, there are numerous methods for the generation of OAM, as shown in Table 2. However, not all generation methods are suitable for the mm-wave band, such as patch antennas. Patch antennas that can work in the mm-wave band are significantly compact, but the antenna size is too small, which will reduce antenna gain and not allow for the transmission of EM waves to be used. There are five main ways to generate OAM in the mm-wave band: an SPP, metasurfaces, an antenna array, a lens antenna, and transmission grating.
Table 2. The classical OAM generation methods and characteristics.
Uniform circular array (UCA) is a traditional method for generating OAM beams and was originally proposed by Thidé et al. [14], who used a circle with eight dipole antennas, a radius of ${\lambda}$, and a concentric circle with 16 dipole antennas and a radius of ${2}{\lambda}$ to generate OAM waves in different modes. The number of array elements N and the number of OAM modes satisfy the ${-}{N}{/}{2}\,{\lt}\,{l}\,{\lt}\,{N}{/}{2}$ relation. The three-dimensional radiation direction diagram of the OAM generated by the array is shown in Figure 4. An observation can be made that vortex EM waves were generated with modes ${l} = {0}$, 1, 2, and 4 with the toroidal array. The beam exhibited a diverging phase wavefront, and the divergence angle became larger as the number of OAM modes increased. Such a phenomenon will trigger the reduction of antenna gain. In addition, the application of vortex EM waves to long-range communication requires a large aperture or distributed antenna at the receiving end to complete the reception of the signal. It is also necessary to add filtering devices to detect and distinguish different modes of OAM, adding difficulty to the design of OAM transceiver systems. As such, the modal value of the OAM antenna cannot be increased infinitely with the number of elements, and a balance between the quality of the transceiver signal and the channel capacity needs to be maintained.
Figure 4. The OAM 3D radiation direction map. (Source: Thidé et al. [14]; used with permission.)
The principle of UCA is to place the N array elements of antennas in an isotropic and equally spaced manner on a circumference. An equal-spoke excitation with continuous and phase difference ${(}{2}{\pi}{l}{)}{/}{N}$ was applied to the array elements, and the excitation phase of the nth element was [20] \[{\varphi}_{n} = \frac{\left({{n}{-}{1}}\right){2}{\pi}{l}}{N},\quad{n} = {1},{2}\ldots\ldots{.,}{N} \tag{12} \] where l is the OAM mode to be excited and N is the total number of array elements.
A common mm-wave ring array antenna uses a phase shifter to realize the phase delay of antenna array elements [21]. As shown in Figure 5, the authors used eight tunable delay lines and a probe receiver designed to make the measurement more accurate. The collected beam is then sent to an oscilloscope and compared with the reference signal. Therefore, the relative phase delays among all antennas could be measured. Both the number of antennas and the distance from the antenna to the center of the array had an impact on the beam generation and beam quality. Other methods usually rely on the feed network. Wu et al. later proposed a ${2}\,{\times}\,{2}$ broadband bidirectional dual circularly polarized (CP) OAM antenna array at mm-wave frequencies based on the principle of circular phased array [22], which incorporated the circular polarization characteristics of OAM. The antenna array can generate dual-mode OAM simultaneously by sequentially rotating the feed, with the antenna consisting of four bidirectionally radiating dual CP elements and each CP element being rotated by 90° in a clockwise direction sequentially. The antenna cell structure consists of two supersurface patch layers loaded on a bidirectional back-cavity slot antenna based on substrate-integrated waveguide technology. Such structures can generate OAM beams with left-hand circular polarization (LHCP) ${l} = + {1}$ and right-hand circular polarization (RHCP) ${l} = {-}{1}$, with an impedance bandwidth of 26 GHz to 36 GHz and an axial ratio bandwidth of 32.3%.
Figure 5. (a) The tunable delay line array and (b) the arrangement of antennas. (c) The approach for tuning and measuring the relative phase delays among different branches. TDL: tunable delay line. (Source: Xie et al. [21]; used with permission.)
The existing methods for generating mm-wave OAM based on ring arrays have been explored based on UCA, which requires multiport phase shifter devices to generate multimodal OAM beams, thereby significantly increasing the complexity of communication systems. At present, most of the research on the use of a concentric UCA (CUCA) to generate multimodal OAM is still in the theoretical stage. Although CUCA has better flexibility and is beneficial for multimode multiplexing in mm-wave wireless communication, the antenna structures designed based on such principles usually all require complex feed networks and high system complexity.
Metasurfaces have a particular property, namely, the modulation of phase [23]. Since the phase of EM waves, beam deflection, holographic, vortex wave generation, and phantoms can be controlled, several scholars have combined metasurfaces with a reflective surface in the mm-wave band and proposed a metasurfaces-based reflective antenna to generate OAM beams. Notably, a planar reflective array antenna is an antenna form that combines surface and array antenna characteristics. To allow the feed source’s spherical wave to be converted into a planar wave by phase compensation, such an antenna uses discrete plane-shifted units to approximate the conventional reflector antenna’s continuous phase distribution. To expand further, when the incident wave reaches the reflecting surface array, the reflecting array units on the array reflect the incident wave with different phases using resonance or phase delay.
A high-gain plane wave is generated by superimposing the radiation fields generated by each unit on each other over the array surface. The structure of the reflector array antenna diagram is shown in Figure 6. As the feed horn radiation is a low-gain spherical wave, to generate a high-gain beam, the first step is to offset the spatial phase difference from the feed antenna phase center to each unit of the reflector array. The phase difference required for each array element in the reflection array can be expressed as [24] \[\begin{array}{r}{{\varphi}^{\text{unit}}{(}{x}_{i},{y}_{i}{)} = {k}_{0}\left\lfloor{{d}_{i}{-}\sin{\theta}_{0}\,{\times}\,{(}\cos{\varphi}_{0}\,{\times}\,{x}_{i} + \sin{\varphi}_{0}\,{\times}\,{y}_{i}{)}}\right\rfloor}\end{array} \tag{13} \]
Figure 6. The reflector array antenna structure diagram.
where ${\varphi}^{\text{unit}}{(}{x}_{i},{y}_{i}{)}$ is the phase compensation required for the (i, j)th cell; ${k}_{0}$ is the number of free-space waves; ${d}_{i}$ is the distance from the feed source phase center to each ith cell; and ${\theta}_{0}$, ${\varphi}_{0}$ is the beam angle of the outgoing wave number. The reflector array antenna can produce a high-gain plane wave when each cell has the phase compensation as given in the equation. In order for the reflector array antenna to generate effective OAM EM waves, it is necessary not only to compensate for the spatial phase difference of the feed beam but also to perform OAM beam assignment on the reflector array. The radiation field of the reflector array antenna after considering the compensation of the spatial phase difference of the feed source is a high-gain pencil-shaped beam. The main lobe of such a pencil-shaped beam can be regarded as a Gaussian beam with a small width. As previously shown, the OAM EM wave can be obtained by attaching an additional $\exp{(}{-}{il}{\theta}{)}$ phase to the Gaussian beam. By combining the phase compensation required for each cell of the previous reflector array and attaching an additional phase $\exp{(}{-}{il}{\theta}{)}$ to the phase-compensated cells, the phase distribution of the entire OAM reflector array is \begin{align*}{\varphi}^{\text{unit}}\left({{x}_{i},{y}_{i}}\right) & = {k}_{0}\left\lfloor{{d}_{i}{-}\sin{\theta}_{0}\,{\times}\,\left({\cos{\varphi}_{0}\,{\times}\,{x}_{i} + \sin{\varphi}_{0}\,{\times}\,{y}_{i}}\right)}\right\rfloor \\ & \quad+ {l}\,{\times}\,\arctan\left({{y}_{i}{x}_{i}}\right) \tag{14} \end{align*}
From (14), the OAM reflective array antenna array was also around the center of the antenna spiral distribution. The coverage of the same with the usual reflective array was over 360°, which required the reflective array antenna unit to have a significant phase shift range.
The emergence of metal reflective metasurfaces provides a new direction for OAM applications in the mm-wave band [25]. Geometric metasurfaces primarily follow the physical mechanism of Pancharatnam–Berry (PB) phases, but there are two major inherent restrictions. The first restriction is conjugated symmetry, which would exhibit equal and opposite phase profiles under LHCP and RHCP illumination. On the other hand, the generation of the PB phase is accompanied with a space-variant conversion of polarization states following the path on the Poincaré sphere, leading to PB-phase-based metasurfaces that can only phase-modulate a cross-polarized output field. Yuan et al. [26] have presented a general strategy to simultaneously and independently manipulate both copolarized and cross-polarized transmitted outputs under CP incidence. This can be a good alternative to address the inherent limitation of PB-phase-based metasurfaces that only operate in a cross-polarization manner [26]. However, the difficulties of metasurface antenna fabrication still exist. The emergence of 3D printing technology has effectively solved the problem and greatly simplified the difficulty of making antennas [27], as shown in Figure 7. To achieve high efficiency and high bandwidth performance of the antenna, Lin et al. [28] designed a metasurface OAM antenna composed of a high-refractive-index dielectric resonator unit. By properly adjusting the size and rotation angle of the dielectric resonator, broadband and efficient full-phase control were achieved [28]. In addition, a method of employing two orthogonally polarized metallic gratings for forming a Fabry–Perot cavity and incorporating a subwavelength metallic double-split-ring resonator at the center of each unit cell has been demonstrated to eliminate the coupling between the amplitude and phase, thereby allowing for an ultrawideband OAM antenna to be designed [29]. In 2021, Zhang et al. presented a relatively simple method, based on detour phase, to simultaneously and independently generate TE- and TM-polarized vortex beams carrying OAM modes [30]. The antenna is capable of producing OAM beams in both modes with high modal purity in the operating band, as shown in Figure 8. This method provides new ideas for the application of mm-wave dual-polarization communication systems.
Figure 7. The 3D printing model diagram. (Source: Li et al. [27]; used with permission.)
Figure 8. (a) A schematic of the phase gradient metasurface for dual-polarized vortex beam generation. (b) The unit cell for TE polarization. (Source: Zhang et al. [30]; used with permission.)
The method of generating OAM with reflective metasurface antennas is less costly, and the system is relatively simple, which also facilitates the generation of OAM at high-frequency bands. Despite such advantages, such a method is less flexible and usually can only generate a single-mode OAM beam at a single frequency point, which is not conducive to OAM multiplexing research.
The SPP is the most primitive device for generating OAM beams. The structure of the SPP is as shown, with one side being smooth and the other side having a thickness that varies with the angle. The phase term of the azimuthal angle was introduced, and the phase delay generated by the incident wave through the SPP is \[\Delta{\theta}{(}{\theta},{\lambda}{)} = \frac{{2}{\pi}}{\lambda}\left[{\frac{\left({{n}{-}{n}_{0}}\right){h}_{s}{\theta}}{{2}{\pi}} + {nh}_{0}}\right] \tag{15} \]
The modal number l expression is given by \[{l} = {h}_{s}\left({{n}{-}{n}_{0}}\right){\lambda} \tag{16} \] where n is the refractive index of the SPP; ${n}_{0}$ is the refractive index of the surrounding medium; ${\lambda}$ is the wavelength of the incident wave; ${h}_{s}$ is the step height of the SPP; ${h}_{0}$ is the thickness of the substrate of the SPP; and h is the thickness of the SPP, ${h} = {h}_{0} + {h}_{s}{(}{\theta}{/}{2}{\pi}{)}$. According to the SPP phase delay formula, when the rotation angle ${\theta}$ increases from 0 to ${2}{\pi}$, the increment of $\Delta{\theta}$ is ${2}{\pi}{l}$.
The ideal SPP, which has a continuous phase change, increases linearly with the increase of rotation angle ${\theta}$, but most of the SPP (multilevel step SPP) is used in practice due to the limitation of process technology. As shown in Figure 9, the phase change was no longer continuous, and the thickness exhibited a linear increase in a step shape.
Figure 9. An ideal SPP.
As shown in Figure 10, to generate higher-order vortex beams, a stepped surface is usually used instead of a spiral surface to form a stepped SPP [31]. Recently, a modified approach to SPP, namely, the planar (flat) phase plate (FPP) [32], has also been proposed to generate OAM beams. This method differs from the conventional SPP that uses the thickness difference of the phase plate to change the phase of the beam, where the FPP uses different borehole densities or radii to control the change of the dielectric constant to adjust the wavefront of the EM wave. Gaddam et al. [33] also designed an OAM antenna based on an FPP capable of steadily generating OAM beams with ${l} = {1}$, 2, 3, and 4. The desired OAM modulation was generated using horn-excited FPPs. The notable performance of such an antenna for high data rate transmission was verified [33]. This FPP differs from conventional SPPs in that a high-dielectric-constant material with a dielectric constant $\varepsilon\approx{10}{.}{8}$ was used for fabrication. Although the structure of this device is also planar, the structural accuracy is considerably demanding, rendering fabrication of the phase plates more challenging. Thus, Hui et al. [34] applied 3D printing technology to fabricate an ultralow-reflectivity SPP with characteristic main lobes of approximately 10.5° in direction, 14.6 dB in gain, and 14.5° in 3-db beamwidth, as shown in Figure 11. Compared with the SPP without impedance matching, the reflectivity was improved by over 20 dB. Such impedance matching can significantly reduce the reflections when SPPs are made with high-dielectric-constant materials [34]. The cost and system complexity of the method of applying SPP to generate OAM are relatively low. However, SPP systems are usually less flexible and can only generate a single OAM beam.
Figure 10. A stepped spiral surface. (Source: Zhu et al. [31]; used with permission.)
Figure 11. FPPs for horn excitation [34]. (a) Stereoscopic lens. (b) Planar lens.
As shown in Figure 12, lens antennas can be divided into stereoscopic lenses led by lobular lenses and columnar lenses [35], and planar lenses based on metamaterials [36]. Among the variety of lens antennas, research on the stereo lens began earlier and is more mature with better focusing characteristics. However, due to the large sizes of such lenses, there are difficulties in processing the characteristics of large-scale use. The processing and manufacturing of planar lenses is relatively simple, and the flat structure of such lenses is easier to integrate into microwave circuits. The planar lens is based on artificial EM metamaterials, that is, the transmission spiral structure. By introducing different phase shift values in different regions, an EM transmission wave with phase differences varying with the region is generated. Such a method has low cost and low system complexity and is a notable solution for realizing vortex beams. In most existing studies, an ordinary directional horn antenna was used as the feed source of the lens, which gives different phase shift values at different directional angles on the surface of the plane lens, forming a spiral phase surface. When ${l} = {1}$, the phase delay for one week of rotation around the surface of the plane lens is exactly 360°; when ${l} = {2}$, the phase delay for one week of rotation around the surface of the plane lens is precisely 720°; and so on. To simplify the communication system, Bai et al. [37] integrated a conventional horn antenna with a single perforated planar dielectric lens in designing a horn lens antenna operating between 45 GHz and 54 GHz, as shown in Figure 13. Such an antenna forms a vortex by varying the diameter of the aperture and obtaining the corresponding azimuthal phase delay [37]. In 2021, Wu et al. [38] introduced broadband characteristics into the mm-wave OAM antenna. A broadband C-type polarization conversion transmitting array element and broadband magnetoelectric (ME) dipole antenna was proposed in which the planar lens’s building block and feed source were used, as shown in Figure 14. The ME dipole does not carry OAM modes. The flat lens can simultaneously convert the spherical wavefronts from two ME dipole sources into transmitted vortex waves with different OAM numbers ${l} = {-}{1}$ and ${l} = + {1}$. At the same time, the experimental results verified that the designed flat-lens OAM antenna has good potential in mm-wave high-speed wireless communication [38].
Figure 12. (a) A stereoscopic lens. (b) A planar lens. (Sources: Lafond et al. [35] and Rahmati and Hassani [36]; used with permission.)
Figure 13. A horn lens antenna. Horn lens antenna. (a) without lens and (b) with lens. (Source: Bai et al. [37]; used with permission.)
Figure 14. A plane lens. (a) 3-D perspective view of the flat lens pixel element. (b) C-shaped metallic pattern in the middle layer. (Source: Wu et al. [38]; used with permission.)
The transmission grating structure is a traditional method for generating high-frequency band OAM beams. A forked grating is obtained from cross-misalignment generated by interferometric fringes, which is a similar method for generating radio OAM to the optical OAM beam generation method. The structure of the transmission grating is similar to the spiral phase structure, and the holographic plate (HP) is generated by simulations using the computational hologram method, and the EM wave carrying the OAM is generated using such an HP. The actual model of the HP is shown in Figure 15 [39]. The device diagram of the holographic method to generate OAM is shown in Figure 16. The first step in the method was to produce a phase hologram using a computer. The transmitted 60-GHz signal was further localized using a brass aperture with a 5-mm gap in the middle before it was radiated on to the HP or SPP surface. The size of the gap was chosen so that it corresponded to the 60-GHz wavelength. Due to the special spiral phase structure of the vortex beam, an interference fringe distribution pattern was produced when the interference contained crossed dislocation structures, and the number of dislocations was the same as the topological charge value. A particular medium was then used to record such patterns to create a forked grating that could be obtained from the vortex beam, thereby allowing for a phase hologram to be obtained. A dielectric plate of the desired shape was drilled and engraved according to the output data of the computer simulation and used to create the holographic phase surface. In [39], polytetrafluoroethylene was used to fabricate the SPP. The beam was illuminated on the fabricated holographic grating to form a vortex beam with various topological charge values that were needed [39]. As demonstrated in the study, 4 Gb/s uncompressed video could be successfully transmitted over an OAM wireless channel in the 60-GHz band. Such spatial multiplexing based on OAM and holographic beam formation would further increase the system capacity. Spatial light modulator (SLM) is also a notable computational holography method. As a computer-controlled method, SLM is simpler and faster to operate, and the images on the computer are directly loaded, reducing the complex process of making holographic gratings [40]. According to the output of the signal, SLMs can be divided into reflective SLMs and transmissive SLMs [41]. Under the action of the SLM, the interferogram generated by the interaction of vortex beams of various eigenvalues could be obtained. As shown in Figure 17, the device is composed of an OAM beam generation simulator, an optical modulator, an optical telescope, and an imaging system. The vortex beam interferometric hologram simulated by the program is loaded onto the SLM, and the eigenvalues and spiral directions of the vortex beam are identified by observing the number of bifurcations and the direction of the bifurcation dislocations in the center of the phase hologram through the focal plane imaging system. Such a method of OAM beam generation using grating structures is generally still used in fiber optic communication, and the application in wireless communication is yet to be developed. By facilitating OAM beam generation in the mm-wave band, such a method can effectively solve the problem of complex OAM generation in the high-frequency band and has a low fabrication cost. However, communication systems based on such a method are more complex.
Figure 15. An HP model drawing. Holographic plate model drawing. (a) l = 1. (b) l = 3. (c) l = 5. (Source: Mahmouli and Walker [39]; used with permission.)
Figure 16. A 4-Gb/s uncompressed wireless OAM experimental setup. (Source: Mahmouli and Walker [39]; used with permission.)
Figure 17. The reflective SLM OAM communication system.
The detection principle of mm-wave OAM is similar to that of lower-band OAM, but the difference lies in how to “derotate.” The current solution is to reverse the phase rotation of the eddy EM wave at the receiver [42], that is, add a reverse rotation phase factor ${e}^{{-}{jl}{\phi}}$ to the eddy EM wave. The difficulty lies in determining the modal value of the vortex EM wave l. The main focus of existing research on vortex EM wave reception has been on detecting the modal value of the vortex EM wave l. According to existing literature, there are five main detection methods for the OAM, as shown in Table 3.
Table 3. The classical OAM detection methods and characteristics.
In conventional communication systems, EM waves reach the receiving antenna through multiple paths such as LOS, reflection, scattering, and refraction from the transmitting antenna. The communication scenarios can be roughly divided into LOS and non-LOS. In contrast, in vortex EM wave systems, the OAM system performance is related to the propagation path of EM waves and the relative position relationship between the transmitting and receiving ends due to the unique spiral phase structure of vortex EM waves [43]. Through traditional classification, the characteristics of vortex EM waves cannot be described in detail; therefore, the communication scenario of OAM systems is divided into the transceiver-aligned scenario and transceiver-unaligned scenario according to the relative position relationship between the antennas at the transceiver end. The unaligned scenario is divided explicitly into off-axis, and nonparallel transceiver-end scenarios since any misalignment scenario is a specific combination of the two scenarios. However, in practical applications, there is communication node movement, which can lead to a Doppler shift and affect the communication quality. Therefore, we also discuss solutions to generate Doppler shifts at different moving speeds.
Most of the current OAM detection approaches are based on alignment scenarios, as shown in Figure 18. The interferometric method is a notable OAM detection approach, and the basic principle is that a reference beam is introduced to interfere with the vortex wave, with the topological charge of the vortex wave and the position thereof being determined by analyzing the shape and number of interferometric fringes. The schematic diagram of the interferometric detection of the OAM beam is shown in Figure 19, where the devices are aligned with each other. Phase singularities in light beams can be artificially created by several methods, such as cylindrical lenses, SPPs, and synthesized holograms. When a plane wave passes through such a phase singularity generation device, a spiral wavefront character is imprinted on the incident plane wave, and the biprism enables overlap of the beams passing through the phase plate and the plane electron wave. If the wave beam to be measured contains a phase singularity, the interference fringes at that location will show a dislocation or bifurcation. Then, the topological load of the vortex wave can be determined from the number of bifurcations of the interference fringe. Uchida and Tonomura [44] passed a plane wave beam through a helical phase plate made of a stacked graphite film, and a vortex wave beam with a phase singularity was generated. The vortex beam and the interference pattern in the transmission electron microscope showed the position of the phase singularity and the shape of the interference fringes, from which the OAM mode was determined [44]. However, when this method is applied to the detection of multimodal OAM beams, the complexity of the receiving device can increase significantly.
Figure 18. The OAM communication model in an alignment scenario.
Figure 19. An interference method schematic diagram.
When CUCAs are used as OAM transmission antennas, two-array alignment is required [45]. CUCAs can demultiplex multiple radio OAM modes. The whole process of OAM demultiplexing can be viewed as some kind of spectral analysis of the received OAM beams [46]. When the mode number l is different from one of the beam modes, the result of the spectral analysis tends to 0, and vice versa. When more than one mode number exists at the same time, a composite mode detection is required. In 2010, two methods for detecting far-field OAM modes were proposed: the single-point estimation method and the phase gradient method [47]. The single-point estimation method uses an approximation of the OAM in the far field to estimate the OAM from single-point measurements of the vertical and lateral components of the electric field. This method is highly efficient for detecting OAMs with a low number of modes. Compared with the single-point estimation method, the detection of OAM beams using the phase gradient method has more intuitive results and also has a certain degree of detection efficiency, as shown in Figure 20. In 2017, Xie et al. proposed a circular phase gradient method [48] for measuring dual-mode RF–OAM beams based on the conventional phase gradient method [49]. Such a method detects a dual-mode OAM beam by measuring the entire phase gradient around the receiving circle centered on and perpendicular to the beam axis, which can be used to obtain sufficient phase gradient information.
Figure 20. A schematic diagram of the phase gradient method. (Source: Mohammadi et al. [47]; used with permission.)
However, the higher the number of OAM modes, the more severe the beam dispersion. To achieve information transmission over longer distances, a larger aperture receiving array is required to converge and capture. Although converging beams can be formed by using concentrating mirrors or parabolic reflectors, there is still the problem of high complexity of the communication system. To address such a problem, in 2007, Thidé et al. identified that the UCA used for OAM field sampling could be considered a discrete Fourier transform [14]. Therefore, the authors also consider the application of the sampling theorem to the OAM reception problem. In 2016, Hu et al. proposed a partial aperture sampling receiving (PASR) method [51], as shown in Figure 21. Initially, PASR was the method used for optical OAM detection [52]. The receiving end aperture is located on a $1/P$ arc concentric with and perpendicular to the OAM transmission spindle. M sampling antennas are uniformly distributed on the $1/P$ arc with antenna angular spacing of ${2}{\pi}{/}{MP}$. If modal mixing is not desired, any two different modes in the transmitted modal set should satisfy $\mod{(}{|}{l}_{n1}{-}{l}_{n2}{|},{P}{)} = {0}$ and $\mod{(}{|}{l}_{n1}{-}{l}_{n2}{|},{MP}{)}\ne{0}$. Despite having several limitations, PASR can simplify the receiver-side setup and is robust to nonideal OAM beams. In 2019, Feng et al. combined Fourier transform theory and sampling theorem to solve the vortex wave reception problem [53]. The general OAM vortex aperture sampling reception method was summarized, and the original sampling theorem was extended to multiple OAM vortex aperture sampling receptions by analogy. A variable scale aperture sampling reception (VSASR) was proposed, and the VSASR method overcame the limitation of equally spaced OAM modes necessary for existing PASR methods to maintain OAM mode orthogonality. As a result, VSASR has higher OAM mode multiplexing utilization than PASR for the same receive aperture. Vortex wave coherence can also be well received. Although the orthogonality can be somewhat corrupted, there are several other crosstalk OAM modes. However, the main transmitted OAM modes can be retained and distinguished with certain digital signal processing methods.
Figure 21. The PASR scheme with M antennas uniformly distributed on a 1/P. (Source: Hu et al. [51]; used with permission.)
Most existing schemes are designed for transceiver alignment scenarios, where different OAM patterns can be easily distinguished at the receiver end. However, in a real wireless communication scenario, the alignment of the transceiver antennas cannot be easily ensured. If the transceiver antennas are not aligned, unequal transmission distances will generate phase turbulence, rendering difficulties in decomposing the signals of different OAM modes. In the present study, the misalignment scenario was divided into the tilted and off-axis transmission, as shown in Figure 22. The reason for division into the two cases is that any misalignment case is a specific combination of the two. Cano et al. revealed that the BER of OAM systems is highly dependent on the alignment between the transceivers [54]. At the same time, Chen et al. investigated the effect of misalignment scenarios on channel capacity and proposed a beam steering scheme to avoid a sharp drop in performance [55]. To a certain extent, the problem of capacity degradation of communication systems in the case of misalignment was mitigated. Moreover, OAM receive transmissions, especially multimode OAM receive transmissions, are considerably sensitive to the alignment errors of transmitters and receivers. To address such issues, in 2017, Chen et al. proposed a reception method in which phase compensation and signal detection were integrated using channel phase information [56]. Such a reception scheme has a more robust anti-interference capability than the conventional scheme, resulting in better communication system performance. Despite such capability, OAM also has a significant problem, namely, the divergence of the OAM beam. As such, a number of researchers have also proposed various solutions [57]. However, such solutions do not consider how to reduce the effect of OAM divergence angle and improve the signal-to-noise ratio when the transmission distance varies in OAM communication systems. Zheng et al. proposed a solution to the problem of signal-to-noise ratio degradation due to the OAM beam divergence angle when transmitting signals in mobile and coaxial scenarios with conventional fixed UCAs [58]. Concentric circular arrays were considered to be placed at the transmitter and receiver ends to improve the system performance by using antenna selection and receiver diversity for the selection of the most matching transceiver array to transmit signals at different distances.
Figure 22. An OAM communication model in nonaligned scenarios. (a) The tilt OAM system model and (b) the off-axis OAM system model.
Table 3 provides a comparison of the common detection methods for radio OAM in static scenarios. These methods typically require consideration of computational complexity. Comparing the multimodal OAM detection methods, both the interferometric phase detection method and the single-point estimation method have high system complexity and computational complexity. Therefore, these two methods are not commonly used in practical applications. By comparison, the phase gradient method is the more frequently used method for practical OAM detection. It is less complex but comes with a drawback that it can only detect single-mode OAM beams. For the detection of multimode OAM beams, the commonly used method is CUCA. the improved PASR based on CUCA has a better performance to some extent. However, the method still needs further experimental validation. Meanwhile, the inverse SPP method is also widely used for detecting OAM modes in wireless OAM communication. In the particular case of LOS wireless communications, CUCAs have been demonstrated to be more suitable for demultiplexing composite OAM modes.
In practice, the nonalignment scenario also has communication node movements, which can produce Doppler shifts. Hence, the channel estimation issue can be addressed effectively, where the associated Doppler offset influences are estimated. This problem also exists with OAM communication systems. We divide them into low- and medium-speed movement scenarios and high-speed movement scenarios for discussion.
In the case of low- and medium-speed movement, the Doppler shift can be estimated using the phase difference method [59]. The idea of this method is to consider the Doppler shift as a fixed constant value in a short period of time, when the moving speed is not fast enough and the moving distance is short, so the Doppler shift can be considered as a constant quantity. The block diagram of the phase difference estimator is shown in Figure 23, which shows the strong path selector, path tracer, Doppler controller, and Doppler calculator. The input of this module is a channel impulse response estimator, and the output is the estimated Doppler spread value.
Figure 23. A block diagram of the phase difference estimator. (Source: adapted from Hadiansyah et al. [59]; used with permission.)
The method consists of the following steps:
This calculation method is simpler, more accurate, and more effective in the estimation of moderate- or low-speed conditions but less effective when moving too fast.
In high-speed scenarios, the fast change in frequency shift leads to inaccurate estimation. Therefore, the Kalman filter algorithm is introduced to improve the estimation performance. This algorithm is derived from the Wiener filter, which is a time-domain-based algorithm. For a linear system and when the system noise signal conforms to a Gaussian distribution, the algorithm can obtain a recursive minimum mean square error estimate of the system state and converge quickly to track changes quickly. The Kalman filter performs the tracking operation using two basic steps of time update and measurement update, as shown in Figure 24. The overall working of the Kalman filter is a recursive process. The prediction step generates the estimates of the current state variables and their associated uncertainties. Using the upcoming measurement value that has random noise influences, the estimates are updated with the help of a weighted average. The average weight is computed by giving the estimates with the higher probability of occurrence a large weight value [60]. On this basis, the authors also proposed a hybrid interval type 2 fuzzy-assisted Kalman filter for channel estimation. The method is able to model efficiently under high uncertainty and has better performance in fast time varying channel conditions with high node mobility in a system.
Figure 24. A Kalman filter. (Source: Kaur et al. [60]; used with permission.)
The elimination of Doppler shift is necessary in many cases. However, the Doppler shift itself carries information about the direction of motion and velocity that can be exploited. Compared with the traditional Doppler radar, which has a “blind spot” for detecting angularly moving targets, the rotational Doppler effect of the vortex EM wave radar makes it capable of detecting rotating moving targets. Complex motion targets can obtain richer motion information and can effectively detect the rotational speed and acceleration of the target. These will be able to effectively improve the motion compensation capability in radar imaging. Vortex EM wave radar based on the rotational Doppler effect will likely play an important role in the detection and warning of tornadoes, vortexes, and other natural disasters. It also has a wide range of application prospects in military fields such as true and false warhead prediction of missiles, wartime environmental situational awareness, and radar countermeasures. This article does not discuss these potential uses.
The emergence of OAM has solved the problem of scarce spectrum resources in wireless communication to some extent. The first OAM-based wireless communication experiment found that OAM can increase wireless communication transmission capacity without increasing bandwidth [61]. This experiment successfully separated two wireless signals of the same frequency and used two sets of antennas to achieve 442 m of planar and vortex EM wave transmission at the San Marco experimental site, respectively. Subsequently, OAM has been increasingly explored in the field of RF. Recent studies have shown that OAM communication systems in the mm-wave band perform better. A 4Gb/s uncompressed video transmission link has been achieved on the 60-GHz OAM radio channel [39]. At the same time, there is an LOS link, and OAM can be used to encode information to guarantee the correct transmitter–receiver alignment [63], which can also be combined with other multiplexing techniques. In another previous study [64], multiple pairs of antennas were used to combine OAM multiplexing with the conventional air division multiplexing technique. Figure 25 shows the schematic of the experimental setup. The signal at 28 GHz is amplified and split into an in-phase and quadrature (I/Q) local oscillation signal, which generates two data streams at 28 GHz, each divided into two paths. Due to the mutual delay, the four data streams are independent of each other. The waveform amplitude is then adjusted to convert it into an OAM beam. The generated beam is passed through two apertures and a power divider and handed over to the MIMO system for processing. Experimental results show that the device achieves a transmission rate of 16 Gb/s by applying a ${2}\,{\times}\,{2}$ antenna aperture structure at the 28-GHz band. High-frequency carriers help reduce transmission loss in the OAM beam and improve communication efficiency. In a related study, an experiment in the 28-GHz band achieved a capacity of 32 Gb/s and a spectral efficiency of about 16 b/s/Hz by combining four OAM modes each carrying 4 Gb/s 16-QAM modulated signals with two polarization states [65].
Figure 25. The experimental setup of a 16-Gb/s mm-wave link using MIMO processing of two OAM beams. LO: local oscillator; AMP: amplifier; AWG: arbitrary waveform generator; Ch: channel; I: in-phase; Q: quadrature. (Source: Ren et al. [64]; used with permission.)
It is well known that MIMO communication systems are of some practical significance for channel capacity enhancement. Therefore, there is some academic controversy about whether OAM can provide new physical dimensions and degrees of freedom. The fundamental reason is that multimodal OAM vortex beams can be generated by array antennas in the form of CUCA, for example, and have similar hardware structures as multiantenna-based MIMO transmission systems, so such OAM vortex beam transmission is often considered as a special case of multiantenna MIMO transmission systems. That is, no new dimensions are generated compared with conventional multiantenna MIMO systems [66]. Of course, in addition to CUCA antennas generating statistical-state OAM vortex beams, some OAM-specific antennas can also generate such vortex beams, such as using SPPs and diffraction gratings. However, in practice, there are differences between the two approaches. First, unlike conventional MIMO systems, OAM receivers are less complex because the inherent orthogonality between OAM modes mitigates interchannel interference. Second, combining OAM with traditional MIMO technology can result in higher capacity gains [67] or provide a more flexible system design [68]. Figure 26 shows the multipath transmission model of the system. The experimental results show that the radius of the antenna array decreases as the reflection distance increases. Moreover, the performance of the OAM–MIMO system in terms of capacity is better than that of the conventional MIMO system. Finally, considering the spatial distribution characteristics of spatially modulated beams, Ge et al. [50] proposed a new OAM spatially modulated mm-wave communication system for future mobile networks. The maximum energy efficiency of the system is improved by 227.2% compared with the OAM–MIMO mm-wave communication system. The system also has superior performance in terms of interference immunity and is suitable for long-distance transmission [50]. The OAM–MIMO system architecture summarized above can be used not only in the field of wireless communication but also in free-space optical communication links [62].
Figure 26. The multipath propagation model. (a) The top view and (b) the side view. CR: ground to the ceiling; CW: ceiling to the walls; CG: ceiling to the ground; GW: ground to the walls; GC: ground to the ceiling; GR: ground receiver; WR: walls receiver; TW: walls transmitter; WC: walls to the ceiling; WG: walls to the ground. (Source: He et al. [67]; used with permission.)
OAM has received considerable attention as a new technology. However, research on the technology in optics is more mature, and further exploration in wireless communication is needed. The methods for generating and detecting OAM beams and the application of OAM in the mm-wave band were reviewed in the present article. In order to further investigate the prospect of OAM in wireless communication, the following issues can be considered in combination with the premise of the present article:
Table 4 lists the abbreviations used in this article.
Table 4. Abbreviations.
This work was supported by the National Natural Science Foundation of China under Grant 62061039. The corresponding author is Xuehong Sun.
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Digital Object Identifier 10.1109/MMM.2023.3269619